Vibrations Fundamentals and Practice ch09

Vibrations Fundamentals and Practice ch09 Maintaining the outstanding features and practical approach that led the bestselling first edition to become a standard textbook in engineering classrooms worldwide, Clarence de Silva''s Vibration: Fundamentals and Practice, Second Edition remains a solid instructional tool for modeling, analyzing, simulating, measuring, monitoring, testing, controlling, and designing for vibration in engineering systems. It condenses the author''s distinguished and extensive experience into an easy-to-use, highly practical text that prepares students for real problems in a variety of engineering fields.

de Silva, Clarence W “Signal Conditioning and Modification”

Vibration: Fundamentals and Practice

Clarence W de Silva

Boca Raton: CRC Press LLC, 2000

9 Signal Conditioning and

voltage-to-fre-9.1 AMPLIFIERS

The level of an electrical signal can be represented by variables such as voltage, current, andpower Across variables, through variables, and power variables that are analogous can be definedfor other types of signals (e.g., mechanical) as well Signal levels at various interface locations

of components in a vibratory system must be properly adjusted for correct performance of thesecomponents and the overall system For example, input to an actuator should possess adequatepower to drive the actuator A signal should maintain its signal level above some threshold duringtransmission so that errors due to signal weakening are not excessive Signals applied to digitaldevices must remain within the specified logic levels Many types of sensors produce weak signalsthat must be upgraded before they can be fed into a monitoring system, data processor, controller,

or data logger

Signal amplification concerns proper adjustment of the signal level for performing a specifictask Amplifiers are used to accomplish signal amplification An amplifier is an active device thatneeds an external power source to operate Although active circuits — amplifiers in particular —can be developed in the monolithic form using an original integrated-circuit (IC) layout so as toaccomplish a particular amplification task, it is convenient to study their performance using the

operational amplifier (op-amp) as the basic building block Of course, operational amplifiers arewidely used not only for modeling and analyzing other types of amplifiers, but also as basicbuilding blocks in building these various kinds of amplifiers For these reasons, the present

discussion on amplifiers will focus on the operational amplifier An introduction to this topic waspresented in Chapter 8.

9.1.1 O PERATIONAL A MPLIFIER

The origin of the operational amplifier dates back to the 1940s when the vacuum tube operationalamplifier was introduced Operational amplifier or op-amp got its name due to the fact that it wasoriginally used almost exclusively to perform mathematical operations; for example, in analogcomputers Subsequently, in the 1950s, the transistorized op-amp was developed It used discreteelements such as bipolar junction transistors and resistors Still, it was too large in size, consumedtoo much power, and was too expensive for widespread use in general applications This situationchanged in the late 1960s when the integrated-circuit (IC) op-amp was developed in the monolithicform, as a single IC chip Today, the IC op-amp, which consists of a large number of circuit elements

on a substrate of typically a single silicon crystal (the monolithic form), is a valuable component

in almost any signal modification device

An op-amp can be manufactured in the discrete-element form using, say, ten bipolar junctiontransistors and as many discrete resistors or alternatively (and preferably) in the modern monolithicform as an IC chip that may be equivalent to over 100 discrete elements In any form, the devicehas an input impedance Z i, an output impedance Z o, and a gain K Hence, a schematic model for

an op-amp can be given as in Figure 9.1(a) The conventional symbol of an op-amp is shown in

Figure 9.1(b) Typically, there are about six terminals (lead connections) to an op-amp For example,there are two input leads (a positive lead with voltage v ip and a negative load with voltage v in), anoutput lead (voltage v o), two bipolar power supply leads (+v s and –v s), and a ground lead.Note from Figure 9.1(a) that under open-loop (no feedback) conditions,

v in = 0), then

(9.3)

and if the positive input lead is grounded (i.e., v ip = 0), then

(9.4)Accordingly, v ip is termed noninverting input and v in is termed inverting input

v o =Kv i

v i=v ip−v in

v o=Kv ip

v o= −Kv in

E XAMPLE 9.1

Consider an op-amp having an open loop gain of 1 × 105 If the saturation voltage is 15 V, determinethe output voltage in the following cases:

(a) 5 µV at the positive lead and 2 µV at the negative lead

(b) –5 µV at the positive lead and 2 µV at the negative lead

(c) 5 µV at the positive lead and –2 µV at the negative lead

(d) –5 µV at the positive lead and –2 µV at the negative lead

(e) 1 V at the positive lead and negative lead grounded

(f) 1 V at the negative lead and positive lead grounded

Field effect transistors (FETs) — for example, metal oxide semiconductor field effect transistors

(MOSFETs) — could be used in the IC form of an op-amp The MOSFET type has advantages

over many other types; for example, higher input impedance and more stable output (almost equal

to the power supply voltage) at saturation, making the MOSFET op-amps preferable over bipolar

junction transistor op-amps in many applications

In analyzing operational amplifier circuits under unsaturated conditions, one can use the

fol-lowing two characteristics of an op-amp:

1 Voltages of the two input leads should be (almost) equal

2 Currents through each of the two input leads should be (almost) zero

As explained earlier, the first property is credited to high open-loop gain, and the second property

to high input impedance in an operational amplifier These two properties are repeatedly used to

obtain input-output equations for amplifier circuits and systems

9.1.2 U SE OF F EEDBACK IN O P - AMPS

An operational amplifier is a very versatile device, primarily due to its very high input impedance,

low output impedance, and very high gain However, it cannot be used without modification as an

amplifier because it is not very stable in the form shown in Figure 9.1 Two factors that contribute

to this problem are:

1 Frequency response

2 Drift

Stated another way, op-amp gain K does not remain constant; it can vary with the frequency of the

input signal (i.e., frequency response function is not flat in the operating range); and also, it can

vary with time (i.e., drift) Frequency response problems arise due to circuit dynamics of an

operational amplifier This problem is usually not severe unless the device is operated at very high

frequencies Drift problems arise due to the sensitivity of gain K to environmental factors such as

temperature, light, humidity, and vibration, and as a result of the variation of K due to aging Drift

in an op-amp can be significant, and steps should be taken to remove this problem

It is virtually impossible to avoid gain drift and frequency-response error in an operational

amplifier But an ingenious way has been found to remove the effect of these two problems at the

amplifier output Because gain K is very large, by using feedback, one can virtually eliminate its

effect at the amplifier output This closed-loop form of an op-amp is preferred in almost every

application In particular, voltage followers and charge amplifiers are devices that use the properties

TABLE 9.1 Solution to Example 9.1

v ip v in v i v o

5 µ V 2 µ V 3 µ V 0.3 V –5 µ V 2 µ V –7 µ V –0.7 V

5 µ V –2 µ V 7 µ V 0.7 V –5 µ V –2 µ V –3 µ V –0.3 V

1 V 0 1 V 15 V

0 1 V –1 V –15 V

of high Z i, low Z o, and high K of the op-amp, along with feedback through a precision resistor, to

eliminate errors due to non-constant K In summary, an operational amplifier is not very useful in

its open-loop form, particularly because gain K is not steady But because K is very large, the

problem can be removed using feedback It is this closed-loop form that is commonly used in

practical applications of an op-amp

In addition to the nonsteady nature of gain, there are other sources of error that contribute to

the less-than-ideal performance of an operational amplifier circuit Noteworthy are:

1 The offset current present at the input leads due to bias currents that are needed to operate

the solid-state circuitry

2 The offset voltage that might be present at the output even when the input leads are open

3 The unequal gains corresponding to the two input leads (i.e., the inverting gain not equal

to the noninverting gain).

Such problems can produce nonlinear behavior in op-amp circuits, and they can be reduced by

proper circuit design and through the use of compensating circuit elements

9.1.3 V OLTAGE , C URRENT , AND P OWER A MPLIFIERS

Any type of amplifier can be constructed from scratch in the monolithic form as an IC chip, or in

the discrete form as a circuit containing several discrete elements such as discrete bipolar junction

transistors or discrete field effect transistors, discrete diodes, and discrete resistors However, almost

all types of amplifiers can also be built using operational amplifier as the basic building block

Because one is already familiar with op-amps, and because op-amps are extensively used in general

amplifier circuitry, the latter approach — which uses discrete op-amps for the modeling of general

amplifiers — is preferred

If an electronic amplifier performs a voltage amplification function, it is termed a voltage

amplifier These amplifiers are so common that the term amplifier is often used to denote a voltage

amplifier A voltage amplifier can be modeled as

(9.5)where

v o = output voltage

v i = input voltage

K v= voltage gain

Voltage amplifiers are used to achieve voltage compatibility (or level shifting) in circuits

Current amplifiers are used to achieve current compatibility in electronic circuits A current

amplifier can be modeled by

(9.6)where

Note that the voltage follower has K v = 1 and, hence, it can be considered as a current amplifier Also,

it provides impedance compatibility and acts as a buffer between a low-current (high-impedance)output device (the device that provides the signal) and a high-current (low-impedance) input device

(device that receives the signal) that are interconnected Hence, the name buffer amplifier or impedance

transformer is sometimes used for a current amplifier with unity voltage gain.

If the objective of signal amplification is to upgrade the associated power level, then a power

amplifier should be used for that purpose A simple model for a power amplifier is

(9.7)where

a signal path (e.g., sensing, data acquisition, and signal generation) where signal levels and powerlevels are relatively low Power amplifiers are typically used in the final stages (e.g., actuation,recording, display) where high signal levels and power levels are usually required

Figure 9.2(a) shows an op-amp-based voltage amplifier Note the feedback resistor R f that servesthe purposes of stabilizing the op-amp and providing an accurate voltage gain The negative lead

is grounded through an accurately known resistor R To determine the voltage gain, recall that the voltages at the two input leads of an op-amp should be virtually equal The input voltage v i is

applied to the positive lead of the op-amp Then the voltage at point A should also be equal to v i.Next, recall that the current through the input lead of an op-amp is virtually 0 Hence, by writing

the current balance equation for the node point A, one obtains

This gives the amplifier equation

(9.9)

Hence, the voltage gain is given by

(9.10)

Note the K v depends on R and R f, and not on the op-amp gain Hence, the voltage gain can be

accurately determined by selecting the two resistors R and R precisely Also note that the output

o i

f i

voltage has the same sign as the input voltage Hence, this is a noninverting amplifier If the voltages are of the opposite sign, it will be an inverting amplifier.

A current amplifier is shown in Figure 9.2(b) The input current i i is applied to the negative

lead of the op-amp as shown, and the positive lead is grounded There is a feedback resistor R f connected to the negative lead through the load R L The resistor R f provides a path for the input

current because the op-amp takes in virtually zero current There is a second resistor R through

which the output is grounded This resistor is needed for current amplification To analyze the

amplifier, note that the voltage at point A (i.e., at the negative lead) should be 0 because the positive lead of the op-amp is grounded (zero voltage) Furthermore, the entire input current i i passes through

resistor R f as shown Hence, the voltage at point B is R f i i Consequently, current through resistor

R is R f i i /R, which is positive in the direction shown It follows that the output current i o is given by

= +

mentation amplifiers, gain is programmable and can be set by means of digital logic Instrumentation

amplifiers are normally used with low-voltage signals

Differential Amplifier

Usually, an instrumentation amplifier is also a differential amplifier (sometimes termed a difference

amplifier) Note that in a differential amplifier, both input leads are used for signal input, whereas

in a single-ended amplifier, one of the leads is grounded and only one lead is used for signal input

Ground-loop noise can be a serious problem in single-ended amplifiers Ground-loop noise can be

effectively eliminated using a differential amplifier because noise loops are formed with both inputs

of the amplifier and, hence, these noise signals are subtracted at the amplifier output Because thenoise level is almost the same for both inputs, it is canceled Note that any other noise (e.g., 60-Hzline noise) that might enter both inputs with the same intensity will also be canceled out at theoutput of a differential amplifier

A basic differential amplifier that uses a single op-amp is shown in Figure 9.3(a) The output equation for this amplifier can be obtained in the usual manner For example, because current

input-through the op-amp is negligible, current balance at point B gives

o

f i

(9.13)

Two things are clear from equation (9.13) First, the amplifier output is proportional to the “difference”

and not the absolute value of the two inputs v i1 and v i2 Second, the voltage gain of the amplifier

is R f /R This is known as the differential gain Note that the differential gain can be accurately set using high-precision resistors R and R f

The basic differential amplifier, shown in Figure 9.3(a) and discussed above, is an importantcomponent of an instrumentation amplifier In addition, an instrumentation amplifier should possessthe adjustable gain capability Furthermore, it is desirable to have a very high input impedance andvery low output impedance at each input lead An instrumentation amplifier that possesses these

FIGURE 9.3 (a) A basic differential amplifier, and (b) a basic instrumentation amplifier.

i i

= ( 2− 1)

basic requirements is shown in Figure 9.3(b) The amplifier gain can be adjusted using the precisely

variable resistor R2 Impedance requirements are provided by two voltage-follower type amplifiers,

one for each input, as shown The variable resistance δR4 is necessary to compensate for errors

due to unequal common-mode gain First consider this last aspect, and then obtain an equation for

the instrumentation amplifier

Common Mode

The voltage that is “common” to both input leads of a differential amplifier is known as the

common-mode voltage This is equal to the smaller of the two input voltages If the two inputs are equal,

then the common-mode voltage is obviously equal to each one of the two inputs When v i1 = v i2,

ideally, the output voltage v o should be 0 In other words, ideally, common-mode signals are rejected

by a differential amplifier But because operational amplifiers are not ideal and because they usually

do not have exactly identical gains with respect to the two input leads, the output voltage v o will

not be 0 when the two inputs are identical This common-mode error can be compensated for by

providing a variable resistor with fine resolution at one of the two input leads of the differentialamplifier Hence, in Figure 9.3(b), to compensate for the common-mode error (i.e., to achieve asatisfactory level of common-mode rejection), first the two inputs are made equal and then δR4 iscarefully varied until the output voltage level is sufficiently small (minimum) Usually, δR4 that is

required to achieve this compensation is small compared to the nominal feedback resistance R4.

Since ideally δR4 = 0, one can neglect δR4 in the derivation of the instrumentation amplifierequation Now, note from the basic characteristics of an op-amp with no saturation (voltages atthe two input leads have to be almost identical), that in Figure 9.3(b), the voltage at point 2 should

be v i2 and the voltage at point 1 should be v i1 Furthermore, current through each input lead of an

op-amp is negligible Hence, current through the circuit path B → 2 → 1 → A must be the same.

This gives the current continuity equations

in which V A and V B are the voltages at points A and B, respectively Hence, one obtains

Now, by subtracting the second equation from the first, one obtains the equation for the first stage

of the amplifier; thus,

2

1 1

Note that only the resistor R2 is varied to adjust the gain (differential gain) of the amplifier In

Figure 9.3(b), the two input op-amps (the voltage-follower op-amps) do not have to be exactly

identical as long as the resistors R1 and R2 are chosen to be accurate This is so because the

op-amp parameters such as open-loop gain and input impedance do not enter the op-amplifier equations,provided their values are sufficiently high, as noted earlier

9.1.5 A MPLIFIER P ERFORMANCE R ATINGS

The main factors that affect the performance of an amplifier are:

1 Stability

2 Speed of response (bandwidth, slew rate)

3 Unmodeled signals

The significance of some of these factors has already been discussed

The level of stability of an amplifier, in the conventional sense, is governed by the dynamics

of the amplifier circuitry, and can be represented by a time constant But more important

consid-eration for an amplifier is the “parameter variation” due to aging, temperature, and other mental factors Parameter variation is also classified as a stability issue, in the context of devicessuch as amplifiers, because it pertains to the steadiness of the response when the input is maintained

environ-steady Of particular importance is the temperature drift This can be specified as a drift in the

output signal per unity change in temperature (e.g., mV·°C–1)

The speed of response of an amplifier dictates the ability of the amplifier to faithfully respond

to transient inputs Conventional time-domain parameters such as rise time can be used to represent this Alternatively, in the frequency domain, speed of response can be represented by a bandwidth

parameter For example, the frequency range over which the frequency response function is ered constant (flat) can be taken as a measure of bandwidth Because there is some nonlinearity in

consid-any amplifier, bandwidth can depend on the signal level itself Specifically, small-signal bandwidth

refers to the bandwidth that is determined using small input signal amplitudes

Another measure of the speed of response is the slew rate Slew rate is defined as the largest

possible rate of change of the amplifier output for a particular frequency of operation Since for agiven input amplitude, the output amplitude depends on the amplifier gain, slew rate is usuallydefined for unity gain

Ideally, for a linear device, the frequency response function (transfer function) does not depend

on the output amplitude (i.e., the product of the DC gain and the input amplitude) But for a devicethat has a limited slew rate, the bandwidth (or the maximum operating frequency at which outputdistortions can be neglected) will depend on the output amplitude The larger the output amplitude,the smaller the bandwidth for a given slew rate limit

S OLUTION

Clearly, the amplitude of the rate of change of output signal, divided by the amplitude of the outputsignal, yields an estimate of output frequency Consider a sinusoidal output voltage given by

(9.14)The rate of change of output is

Hence, the maximum rate of change of output is 2πfa Since this corresponds to the slew rate when

f is the maximum allowable frequency, one obtains

Unmodeled signals can be a major source of amplifier error Unmodeled signals include:

1 Bias currents

2 Offset signals

3 Common-mode output voltage

4 Internal noise

In analyzing operational amplifiers, it is assumed that the current through the input leads is 0 This

is not strictly true because bias currents for the transistors within the amplifier circuit have to flowthrough these leads As a result, the output signal of the amplifier will deviate slightly from theideal value

Another assumption made in analyzing op-amps is that the voltage is equal at the two inputleads But, in practice, offset currents and offset voltages are present at the input leads, due tominute discrepancies inherent to the internal circuits within an op-amp

1

1 10

158

= 31.8 kHz

Common-Mode Rejection Ratio (CMRR)

Common-mode error in a differential amplifier was discussed earlier Note that, ideally, the common

mode input voltage (the voltage common to both input leads) should have no effect on the output

voltage of a differential amplifier But because a practical amplifier has unbalances in the internalcircuitry (e.g., gain with respect to one input lead is not equal to the gain with respect to the otherinput lead and, furthermore, bias signals are needed for operation of the internal circuitry), there

will be an error voltage at the output that depends on the common-mode input The common-mode

rejection ratio (CMRR) of a differential amplifier is defined as

(9.16)

where

K = gain of the differential amplifier (i.e., differential gain)

v cm = common-mode input voltage (i.e., voltage common to both input leads)

v ocm = common-mode output voltage (i.e., output voltage due to common-mode inputvoltage)

Note that, ideally, v ocm = 0 and CMRR should be infinity It follows that the larger the CMRR, thebetter the differential amplifier performance

The three types of unmodeled signals mentioned above can be considered as noise In addition,there are other types of noise signals that degrade the performance of an amplifier For example,ground-loop noise can enter the output signal Furthermore, stray capacitances and other types ofunmodeled circuit effects can generate internal noise Usually in amplifier analysis, unmodeledsignals (including noise) can be represented by a noise voltage source at one of the input leads.Effects of unmodeled signals can be reduced using suitably connected compensating circuitry,including variable resistors that can be adjusted to eliminate the effect of unmodeled signals at theamplifier output [e.g., see δR4 in Figure 9.3(b)] Some useful information about operational ampli-fiers is summarized in Box 9.1

AC-Coupled Amplifiers

The DC component of a signal can be blocked off by connecting the signal through a capacitor

(Note that the impedance of a capacitor is 1/(jω C) and hence, at zero frequency, there will be an

infinite impedance.) If the input lead of a device has a series capacitor, the input is AC-coupled; and if the output lead has a series capacitor, then the output is AC-coupled Typically, an AC-coupled amplifier has a series capacitor both at the input lead and the output lead Hence, its frequency response function will have a high-pass characteristic; in particular, the DC components will be filtered out Errors due to bias currents and offset signals are negligible for an AC-coupled amplifier Furthermore, in an AC-coupled amplifier, stability problems are not very serious

9.2 ANALOG FILTERS

Unwanted signals can seriously degrade the performance of a vibration monitoring and analysissystem External disturbances, error components in excitations, and noise generated internallywithin system components and instrumentation are such spurious signals A filter is a device thatallows through only the desirable part of a signal, rejecting the unwanted part

CMRR= Kv

v

cm

ocm

In typical applications of acquisition and processing of a vibration signal, the filtering taskwould require allowing through certain frequency components and filtering out certain otherfrequency components in the signal In this context, one can identify four broad categories of filters:

1 Low-pass filters

2 High-pass filters

3 Bandpass filters

4 Band-reject (or notch) filter

The ideal frequency-response characteristic of each of these four types of filters is shown in Figure 9.4.Note that only the magnitude of the frequency response function is shown It is understood, however,

that the phase distortion of the input signal should also be small within the pass band (the allowed

frequency range) Practical filters are less than ideal Their frequency-response functions do notexhibit sharp cutoffs as in Figure 9.4 and, furthermore, some phase distortion will be unavoidable

A special type of bandpass filter widely used in acquisition and monitoring of vibration signals

(e.g., in vibration testing) is the tracking filter This is simply a bandpass filter with a narrow pass

BOX 9.1 Operational Amplifiers

Ideal op-amp properties:

• Infinite open-loop differential gain

• Infinite input impedance

• Zero output impedance

• Infinite bandwidth

• Zero output for zero differential input

Ideal analysis assumptions:

• Voltages at the two input leads are equal

• Current through either input lead is zero

Definitions:

• Open-loop gain =

• Input impedance =

• Output impedance =

• Bandwidth = Frequency range in which the frequency response is flat (gain is constant)

• Input bias current = Average (DC) current through one input lead

• Input offset current = Difference in the two input bias currents

• Differential input voltage = Voltage at one input lead with the other grounded when theoutput voltage is zero

• Common-mode gain =

• Common-mode rejection ratio (CMRR) =

• Slew rate = Speed at which steady output is reached for a step input

Output voltageVoltage difference at input leadswith no feedback

Voltage between an input lead and groundCurrent through that lead (with the other input lead grounded and the output in open circuit)Voltage between output lead and ground in open circuit

Current through that lead with normal input conditions

Output voltage when input leads are at the same voltage

Common input voltageOpen - loop differential gainCommon - mode gain

band that is frequency-tunable The center frequency (mid-value) of the pass band is variable,usually by coupling it to the frequency of a carrier signal In this manner, signals whose frequencyvaries with some basic variable in the system (e.g., rotor speed, frequency of a harmonic excitationsignal, frequency of a sweep oscillator) can be accurately tracked in the presence of noise The

inputs to a tracking filter are the signal that is being tracked and the variable tracking frequency

FIGURE 9.4 Ideal filter characteristics: (a) low-pass filter; (b) high-pass filter; (c) bandpass filter; and

(d) band-reject (notch) filter.

(carrier input) A typical tracking filter that can simultaneously track two signals is schematically

shown in Figure 9.5

Filtering can be achieved using digital filters as well as analog filters Before digital signal

processing became efficient and economical, analog filters were used exclusively for signal filtering,and are still widely used In an analog filter, the signal is passed through an analog circuit Dynamics

of the circuit will be such that the desired signal components will be passed through, and theunwanted signal components will be rejected Earlier versions of analog filters employed discretecircuit elements such as discrete transistors, capacitors, resistors, and even discrete inductors.Because inductors have several shortcomings, such as susceptibility to electromagnetic noise,unknown resistance effects, and large size, they are rarely used today in filter circuits Furthermore,due to well-known advantages of integrated-circuit (IC) devices, analog filters in the form ofmonolithic IC chips are extensively used today in modem applications and are preferred overdiscrete-element filters Digital filters that employ digital signal processing to achieve filtering arealso widely used today

9.2.1 P ASSIVE F ILTERS AND A CTIVE F ILTERS

Passive analog filters employ analog circuits containing passive elements such as resistors andcapacitors (and sometimes inductors) only An external power supply is not needed in a passivefilter Active analog filters employ active elements and components such as transistors and opera-tional amplifiers in addition to passive elements Because external power is needed for the operation

of the active elements and components, an active filter is characterized by the need for an externalpower supply Active filters are widely available in monolithic integrated-circuit (IC) form and areusually preferred over passive filters

Advantages of active filters include:

1 Loading effects are negligible because active filters can provide a very high inputimpedance and very low output impedance

2 They can be used with low-level signals because signal amplification and filtering can

be provided by the same active circuit

3 They are widely available in a low-cost and compact integrated-circuit form

4 They can be easily integrated with digital devices

5 They are less susceptible to noise from electromagnetic interference

Commonly mentioned disadvantages of active filters include:

FIGURE 9.5 Schematic representation of a two-channel tracking filter.

1 They need an external power supply.

2 They are susceptible to “saturation”-type nonlinearity at high signal levels

3 They can introduce many types of internal noise and unmodeled signal errors (offset,bias signals, etc.)

Note that the advantages and disadvantages of passive filters can be directly inferred from thedisadvantages and advantages of active filters, as given above

Number of Poles

Analog filters are dynamic systems and can be represented by transfer functions, assuming lineardynamics The number of poles of a filter is the number of poles in the associated transfer function.This is also equal to the order of the characteristic polynomial of the filter transfer function

(i.e., order of the filter) Note that poles (or eigenvalues) are the roots of the characteristic equation.

The following discussion will show simplified versions of filters, typically consisting of a singlefilter stage The performance of such a basic filter can be improved at the expense of circuitcomplexity (and increased pole count) Only simple discrete-element circuits are shown for passivefilters Simple operational-amplifier circuits are given for active filters Even here, much morecomplex devices are commercially available, but the purpose is to illustrate underlying principlesrather than to provide descriptions and data sheets for commercial filters

9.2.2 L OW -P ASS F ILTERS

The purpose of a low-pass filter is to allow through all signal components below a certain (cutoff)frequency and block off all signal components above that cutoff Analog low-pass fitters are widely

used as anti-aliasing filters in digital signal processing (see Chapter 4) An error known as aliasing

will enter the digitally processed results of a signal if the original signal has frequency components

above half the sampling frequency (half the sampling frequency is called the Nyquist frequency).

Hence, aliasing distortion can be eliminated if the signal is filtered using a low-pass filter with itscutoff set at the Nyquist frequency, prior to sampling and digital processing This is one of numerousapplications of analog low-pass filters Another typical application would be to eliminate high-frequency noise in a measured vibration response

A single-pole, passive low-pass filter circuit is shown in Figure 9.6(a) An active filter sponding to the same low-pass filter is shown in Figure 9.6(b) It can be shown that the two circuitshave identical transfer functions Hence, it might seem that the op-amp in Figure 9.6(b) is redundant.This is not true, however If two passive filter stages, each similar to Figure 9.6(a), are connectedtogether, then the overall transfer function is not equal to the product of the transfer functions ofthe individual stages The reason for this apparent ambiguity is the circuit loading that arises due

corre-to the fact that the input impedance of the second stage is not sufficiently larger than the outputimpedance of the first stage But, if two active filter stages similar to Figure 9.6(b) are connectedtogether, such loading errors will be negligible because the op-amp with feedback (i.e., a voltagefollower) introduces a very high input impedance and very low output impedance, while maintainingthe voltage gain at unity

To obtain the filter equation for Figure 9.6(a), note that since the output is open circuit (zero

load current), the current through capacitor C is equal to the current through resistor R Hence,

where the filter time constant is

(9.18)Now, from equation (9.17), it follows that the filter transfer function is

(9.19)

From this transfer function, it is clear that an analog low-pass filter is essentially a lag circuit (i.e.,

it provides a phase lag)

FIGURE 9.6 A single-pole low-pass filter: (a) a passive filter stage; (b) an active filter stage; and (c) the

frequency response characteristic.

It can be shown that the active filter stage in Figure 9.6(b) has the same input/output equation.First, because the current through an op-amp lead is almost 0, one obtains from the previous analysis

of the passive circuit stage

(i)

in which v A is the voltage at the node point A Now, because the op-amp with feedback resistor is

in fact a voltage follower, one has

(ii)

Next, by combining equations (i) and (ii), one obtains equation (9.19), as required Repeating, a

major advantage of the active filter version is that the resulting loading error is negligible

The frequency-response function corresponding to equation (9.19) is obtained by setting s = jω;

thus,

(9.20)

This gives the response of the filter when a sinusoidal signal of frequency ω is applied Themagnitude G(jω) of the frequency transfer function gives the signal amplification, and the phase

angle ∠G(jω) gives the phase lead of the output signal with respect to the input The magnitude

curve (Bode magnitude curve) is shown in Figure 9.6(c) Note from equation (9.20) that for smallfrequencies (i.e., ω << 1/τ), the magnitude is approximately unity Hence, 1/τ can be consideredthe cutoff frequency ωc:

(9.21)

E XAMPLE 9.3

Show that the cutoff frequency given by equation (9.21) is also the half-power bandwidth for the

low-pass filter Show that for frequencies much larger than this, the filter transfer function on theBode magnitude plane (i.e., log magnitude vs log frequency) can be approximated by a straight

line with slope –20 dB per decade This slope is known as the roll-off rate.

v v

c =1

1/ 211

12

τ ωj + =

or

Hence, the half-power bandwidth is

(9.22)

This is identical to the cutoff frequency given by equation (9.11)

Now, for ω >> 1/τ (i.e., τω >> 1), equation (9.20) can be approximated by

This has the magnitude

On the log scale,

It follows that the log10 (magnitude) vs log10 (frequency) curve is a straight line with a slope of –1

In other words, when frequency increases by a factor of ten (i.e., a decade), the log10 (magnitude) decreases by unity (i.e., by 20 dB) Hence, the roll-off rate is –20 dB per decade These observations are shown in Figure 9.6(c) Note that an amplitude change by a factor of (or power by a factor

of 2) corresponds to 3 dB Hence, when the DC (zero-frequency) magnitude is unity (0 dB), the half-power magnitude is –3dB



The cutoff frequency and the roll-off rate are the two main design specifications for a low-passfilter Ideally, one would like a low-pass filter magnitude curve to be flat up to the required pass-band limit (cutoff frequency) and then roll off very rapidly The low-pass filter shown in Figure9.6 only approximately meets these requirements In particular, the roll-off rate is not large enough.One would like a roll-off rate of at least –40 dB per decade and, preferably, –60 dB per decade inpractical filters This can be realized using a higher-order filter (i.e., a filter having many poles).The low-pass Butterworth filter is a widely used filter of this type

11

12

2 2

τ ω + =

τ ω2 2

1 2+ =

τ ω2 21

=

ωτ

Low-Pass Butterworth Filter

A low-pass Butterworth filter having two poles can provide a roll-off rate of –40 dB per decade,and one having three poles can provide a roll-off rate of –60 dB per decade Furthermore, thesteeper the roll-off slope, the flatter the filter magnitude curve within the pass band A two-pole,low-pass Butterworth filter is shown in Figure 9.7 One can construct a two-pole filter simply byconnecting together two single-pole stages of the type shown in Figure 9.6(b) One would thenrequire two op-amps; whereas the circuit shown in Figure 9.7 achieves the same objective usingonly one op-amp (i.e., at a lower cost)

E XAMPLE 9.4

Show that the opamp circuit in Figure 9.7 is a low-pass filter having two poles What is the transferfunction of the filter? Estimate the cutoff frequency under suitable conditions Show that the roll-offrate is –40 dB per decade

S OLUTION

To obtain the filter equation, one writes the current balance equations Specifically, the sum of

currents through R1 and C1 passes through R2 The same current passes through C2 because current

through the op-amp lead must be 0 Hence,

(i)

Also, because the op-amp with a feedback resistor R f is a voltage follower (with unity gain), oneobtains

(ii) From equations (i) and (ii),

2 2

i− A + o − A = o

1

Now, defining the constants

(9.23)(9.24)(9.25)

and introducing the Laplace variable s, one can eliminate v A by substituting equation (iv) into (iii);

thus,

(9.26)

This second-order transfer function becomes oscillatory if (τ2 + τ3)2 < 4τ1τ2 Ideally, one wouldlike to have a zero resonant frequency, which corresponds to a damping ratio value Sincethe undamped natural frequency is

A− o = o

2 2

τ1=R C1 1τ2=R C2 2τ3=R C1 2

ωr = 1 2− ζ ω2 n

τ2 τ3 τ τ2

1 22+

On a log (magnitude) vs log (frequency) scale, this function is a straight line with slope = –2.Hence, when the frequency increases by a factor of 10 (i.e., one decade), the log10 (magnitude)drops by 2 units (i.e., 40 dB) In other words, the roll-off rate is –40 dB per decade Also, ωn can

be taken as the filter cutoff frequency Hence,

9.2.3 H IGH -P ASS F ILTERS

Ideally, a high-pass filter allows through it all signal components above a certain (cutoff) frequency,and blocks off all signal components below that frequency A single-pole, high-pass filter is shown

in Figure 9.8 As for the low-pass filter discussed earlier, the passive filter stage [Figure 9.8(a)]and the active filter stage [Figure 9.8(b)] have identical transfer functions The active filter is desired,however, because of its many advantages, including negligible loading error due to high inputimpedance and low output impedance of the op-amp voltage follower that is present in this circuit.The filter equation is obtained by considering the current balance in Figure 9.8(a), noting thatthe output is in open circuit (zero load current) Accordingly,

o o

o o i

Note that this corresponds to a lead circuit (i.e., an overall phase lead is provided by this transfer

function) The frequency response function is

FIGURE 9.8 A single-pole, high-pass filter: (a) a passive filter stage; (b) an active filter stage; and the

(c) frequency response characteristic.

c =1

pass filter is equal to the cutoff frequency given by equation (9.37), as in the case of the basic pass filter The roll-up slope of the single-pole, high-pass filter is 20 dB per decade Steeper slopesare desirable Multiple-pole, high-pass Butterworth filters can be constructed to give steeper roll-

low-up slopes and reasonably flat pass-band magnitude characteristics

To obtain the filter equation, first consider the high-pass portion of the circuit shown in

Figure 9.9(a) Since the output is open-circuit (zero current), from equation (9.35), one obtains

(9.40)

Now, on eliminating v A by substituting equation (i) into (iii), one obtains the bandpass filter transfer

function:

v v

s s

o

A

=+

One can show that the roots of the characteristic equation

(9.42)are real and negatives The two roots are denoted by –ωc1 and –ωc2, and they provide the two cutofffrequencies shown in Figure 9.9(c) It can be verified that, for this basic bandpass filter, the roll-upslope is +20 dB per decade, and the roll-down slope is –20 dB per decade These slopes are notsufficient in many applications Furthermore, the flatness of the frequency response within the passband of the basic filter is also not adequate More complex (higher-order) bandpass filters withsharper cutoffs and flatter pass bands are commercially available

FIGURE 9.9 Bandpass filter: (a) a basic passive filter stage; (b) a basic active filter stage; and the (c) frequency

τ τ1 2 τ τ τ2

Resonance-Type Bandpass Filters

There are many applications where a filter with a very narrow pass band is required The trackingfilter mentioned in the beginning of this section on analog filters is one such application A filter

circuit with a sharp resonance can serve as a narrow-band filter Note that the cascaded RC circuit

shown in Figure 9.9 does not provide an oscillatory response (filter poles are all real) and, hence,

it does not form a resonance-type filter A slight modification to this circuit using an additional

resistor R1 as shown in Figure 9.10(a) will produce the desired effect

To obtain the filter equation, note that for the voltage follower unit,

s s

A

B

=+

(τ2τ2 1)

Finally, current balance at node B gives

or, using the Laplace variable, one obtains

(iii) Now, by eliminating v A and v B in equations (i) through (iii), the filter transfer function is obtained as

(9.43)

It can be shown that, unlike equation (9.41), the present characteristic equation

(9.44)can possess complex roots

E XAMPLE 9.5

Verify that the bandpass filter shown in Figure 9.10(a) can have a frequency response with a resonantpeak as shown in Figure 9.10(b) Verify that the half-power bandwidth ∆ω of the filter is given by2ζωr at low damping values (Note: ζ = damping ratio and ω r = resonant frequency.)

which has the roots

which are obviously complex

To obtain an expression for the half-power bandwidth of the filter, note that the filter transferfunction can be written as

τ τ1 2 τ τ τ2

For low damping, resonant frequency ωr≅ωn The corresponding peak magnitude M is obtained

by substituting ω = ωn in equation (9.46) and taking the transfer function magnitude; thus,

1 2

2 2

2 2

Accordingly, by solving these two quadratic equations and selecting the appropriate sign, one obtains

(9.49)

(9.50)

The half-power bandwidth is

(9.51)Now, since ωn ≅ ωr for low ζ, one has

(9.52)



A notable shortcoming of a resonance-type filter is that the frequency response within the bandwidth(pass band) is not flat Hence, quite nonuniform signal attenuation takes place inside the pass band

9.2.5 B AND -R EJECT F ILTERS

Band-reject filters, or notch filters, are commonly used to filter out a narrow band of noise

components from a signal For example, 60-Hz line noise in signals can be eliminated by using anotch filter with a notch frequency of 60 Hz

An active circuit that could serve as a notch filter is shown in Figure 9.11(a) This is known

as the Twin T circuit because its geometric configuration resembles two T-shaped circuits connected

together To obtain the filter equation, note that the voltage at point P is v o because of unity gain

of the voltage follower Now, one can write the current balance at nodes A and B; thus,

Next, since the current through the + lead of the op-amp (voltage follower) is 0, the current

continuity through point P is given by

These three equations are written in the Laplace form as

This is known as the notch frequency The magnitude of the frequency-response function of the

notch filter is sketched in Figure 9.11(b) One notices that any signal component at frequency ωo

will be completely eliminated by the notch filter Sharp roll-down and roll-up are needed to allowthe other (desirable) signal components through without too much attenuation

Whereas the previous three types of filters achieve their frequency-response characteristicsthrough the poles of the filter transfer function, a notch filter achieves its frequency-responsecharacteristic through its zeros (roots of the numerator polynomial equation) Some useful infor-mation about filters is summarized in Box 9.2

9.3 MODULATORS AND DEMODULATORS

Sometimes, signals are deliberately modified to maintain the accuracy during signal transmission,

conditioning, and processing In signal modulation, the data signal, known as the modulating signal,

is used to vary a property (such as amplitude or frequency) of a carrier signal Then, one can say

that the carrier signal is modulated by the data signal After transmitting or conditioning themodulated signal, the data signal is usually recovered by removing the carrier signal This is known

as demodulation or discrimination.

Many modulation techniques exist, and several other types of signal modification (e.g., digitizing)can be classified as signal modulation although they might not be commonly termed as such Fourtypes of modulation are illustrated in Figure 9.12 In amplitude modulation (AM), the amplitude of

a periodic carrier signal is varied according to the amplitude of the data signal (modulating signal),

the frequency of the carrier signal (carrier frequency) being kept constant Suppose that the transient

signal shown in Figure 9.12(a) is used as the modulating signal A high-frequency sinusoidal signal

is used as the carrier signal The resulting amplitude-modulated signal is shown in Figure 9.12(b).Amplitude modulation is used in telecommunications, radio and TV signal transmission, instrumen-

o=1

tation, and signal conditioning The underlying principle is useful in other applications such as faultdetection and diagnosis in rotating machinery.

In frequency modulation (FM), the frequency of the carrier signal is varied in proportion tothe amplitude of the data signal (modulating signal), while keeping the amplitude of the carriersignal constant If the data signal shown in Figure 9.12(a) is used to frequency-modulate a sinusoidalcarrier signal, then the result will appear as in Figure 9.12(c) Because information is carried asfrequency rather than amplitude, any noise that might alter the signal amplitude will have virtually

no effect on the transmitted data Hence, FM is less susceptible to noise than AM Furthermore,since in FM the carrier amplitude is kept constant, signal weakening and noise effects that areunavoidable in long-distance data communication will have less effect than in the case of AM,particularly if the data signal level is low in the beginning But more sophisticated techniques andhardware are needed for signal recovery (demodulation) in FM transmission because FM demod-ulation involves frequency discrimination rather than amplitude detection Frequency modulation

is also widely used in radio transmission and in data recording and replay

In pulse-width modulation (PWM), the carrier signal is a pulse sequence The pulse width is

changed in proportion to the amplitude of the data signal, while keeping the pulse spacing constant.This is illustrated in Figure 9.12(d) Pulse-width-modulated signals are extensively used in controllingelectric motors and other mechanical devices such as valves (hydraulic, pneumatic) and machine

FIGURE 9.11 A notch filter: (a) an active twin T filter circuit, and the (b) frequency response characteristic.

tools Note that in a given (short) time interval, the average value of the pulse-width-modulatedsignal is an estimate of the average value of the data signal in that period Hence, PWM signals can

be used directly in controlling a process, without having to demodulate it Advantages of width modulation include better energy efficiency (less dissipation) and better performance withnonlinear devices For example, a device may stick at low speeds due to Coulomb friction This can

pulse-be avoided by using a PWM signal that provides the signal amplitude necessary to overcome friction,while maintaining the required average control signal, which might be very small

In pulse-frequency modulation (PFM) as well, the carrier signal is a pulse sequence In this

method, the frequency of the pulses is changed in proportion to the data signal level, while keepingthe pulse width constant Pulse-frequency modulation has the advantages of ordinary frequencymodulation Additional advantages result due to the fact that electronic circuits (digital circuits inparticular) can handle pulses very efficiently Furthermore, pulse detection is not susceptible tonoise because it involves distinguishing between presence and absence of a pulse rather thanaccurate determination of the pulse amplitude (or width) Pulse-frequency modulation can be used

in place of pulse-width modulation in most applications, with better results

Another type of modulation is phase modulation (PM) In this method, the phase angle of the

carrier signal is varied in proportion to the amplitude of the data signal

Conversion of discrete (sampled) data into the digital (binary code) form is also considered

modulation In fact, this is termed pulse-code modulation (PCM) In this case, each discrete data

sample is represented by a binary number containing a fixed number of binary digits (bits) Since

BOX 9.2 Filters

Active Filters (Need External Power)

Advantages:

• Smaller loading errors (have high input impedance and low output impedance, and hence

do not affect the input circuit conditions and output signals)

• Lower cost

• Better accuracy

Passive Filters (No External Power, Use Passive Elements)

Advantages:

• Usable at very high frequencies (e.g., radio frequency)

• No need for a power supply

• Bandpass: Allows frequency components within an interval and rejects the rest.

• Notch (or band reject): Rejects frequency components within an interval (usually, narrow)

and allows the rest

Definitions

• Filter order: Number of poles in the filter circuit transfer function.

• Anti-aliasing filter: Low-pass filter with cutoff at less than half the sampling rate

(i.e., Nyquist frequency), for digital processing

• Butterworth filter: A high-order filter with a very flat pass band.

• Chebyshev filter: An optimal filter with uniform ripples in the pass band.

• Sallen-Key filter: An active filter whose output is in phase with input.

each digit in the binary number can take only two values, 0 or 1, it can be represented by theabsence or presence of a voltage pulse Hence, each data sample can be transmitted using a set of

pulses This is known as encoding At the receiver, the pulses have to be interpreted (or decoded)

in order to determine the data value As with any other pulse technique, PCM is quite immune tonoise because decoding involves detection of the presence or absence of a pulse rather thandetermination of the exact magnitude of the pulse signal level Also, because pulse amplitude isconstant, long-distance signal transmission (of this digital data) can be accomplished without thedanger of signal weakening and associated distortion Of course, there will be some error introduced

by the digitization process itself, which is governed by the finite word size (or dynamic range) of

FIGURE 9.12 (a) Modulating signal (data signal); (b) amplitude-modulated (AM) signal; (c)

frequency-modulated (FM) signal; (d) pulse-width-frequency-modulated (PWM) signal; and (e) pulse-frequency-frequency-modulated (PFM) signal.

the binary data element This is known as quantization error and is unavoidable in signal

of the data signal, and a negative range of phase angles (say –π to 0) could be assigned for thenegative values of the signal

9.3.1 A MPLITUDE M ODULATION

Amplitude modulation can naturally enter into many physical phenomena More important, perhaps,

is the deliberate (artificial) use of amplitude modulation to facilitate data transmission and signalconditioning First, examine the related mathematics

Amplitude modulation is achieved by multiplying the data signal (modulating signal) x(t) by

a high-frequency (periodic) carrier signal x c (t) Hence, the amplitude-modulated signal x a (t) is given

by

(9.57)

Note that the carrier can be any periodic signal, such as harmonic (sinusoidal), square wave, or

triangular The main requirement is that the fundamental frequency of the carrier signal (carrier

frequency) f c be significantly larger (say, by a factor of 5 or 10) than the highest frequency of

interest (bandwidth) of the data signal Analysis can be simplified by assuming a sinusoidal carrier

frequency; thus,

(9.58)

Modulation Theorem

This is also known as the frequency-shifting theorem and relates the fact that if a signal is multiplied

by a sinusoidal signal, the Fourier spectrum of the product signal is simply the Fourier spectrum

of the original signal shifted through the frequency of the sinusoidal signal In other words, the

Fourier spectrum X a (f) of the amplitude-modulated signal x a (t) can be obtained from the Fourier spectrum X(f) of the data signal x(t) simply by shifting through the carrier frequency f c

To mathematically explain the modulation theorem, one can use the definition of the Fourierintegral transform (see Chapter 4) to obtain

one has

(9.59)

Equation (9.59) is the mathematical statement of the modulation theorem It is illustrated by anexample in Figure 9.13 Consider a transient signal x(t) with a (continuous) Fourier spectrum X(f)

whose magnitude X(f) is as shown in Figure 9.13(a) If this signal is used to amplitude-modulate

a high-frequency sinusoidal signal, the resulting modulated signal x a (t) and the magnitude of its

Fourier spectrum are as shown in Figure 9.13(b) It should be kept in mind that the magnitude has

been multiplied by a c /2 Note that the data signal is assumed to be band limited, with bandwidth f b

Of course, the theorem is not limited to band-limited signals; but for practical reasons, there needs

to be some upper limit on the useful frequency of the data signal Also, for practical reasons (not

for the theorem itself), the carrier frequency f c should be several times larger than f b so that there

is a reasonably wide frequency band from 0 to (f c – f b) within which the magnitude of the modulatedsignal is virtually zero The significance of this should be clear when the applications of amplitudemodulation are discussed

Figure 9.13 shows only the magnitude of the frequency spectra It should be remembered,however, that every Fourier spectrum has a phase-angle spectrum as well This is not shown forconciseness; but, clearly the phase-angle spectrum is also similarly affected (frequency shifted) byamplitude modulation

Side Frequencies and Side Bands

The modulation theorem, as described above, assumed transient data signals with associatedcontinuous Fourier spectra The same ideas are applicable to periodic signals (with discrete spectra)

as well The case of periodic signals is merely a special case of what was discussed above Thiscase can be analyzed using Fourier integral transform itself, from the start In that case, however,one must cope with impulsive spectral lines Alternatively, Fourier series expansion could beemployed to avoid the introduction of impulsive discrete spectra into the analysis But, as shown

in Figure 9.13(c) and (d), no analysis is actually needed for the periodic signal case because thefinal answer can be deduced from the transient signal results Specifically, each frequency compo-

nent f o with amplitude a/2 in the Fourier series expansion of the data signal will be shifted by ± f c

to the two new frequency locations f c + f o and –f c + f o with an associated amplitude aa c/4 The

negative frequency component –f o should also be considered in the same way, as illustrated in

Figure 9.13(d) Note that the modulated signal does not have a spectral component at carrier

frequency f c but, rather, on each side of it, at f c ± fo Hence, these spectral components are termed

side frequencies When a band of side frequencies is present, one has a side band Side frequencies

are very useful in fault detection and diagnosis of rotating machinery

cos2 1 exp exp

9.3.2 A PPLICATION OF A MPLITUDE M ODULATION

The main hardware component of an amplitude modulator is an analog multiplier It is commercially

available in the monolithic IC form, or one can be assembled using integrated-circuit op-amps andother discrete circuit elements A schematic representation of an amplitude modulator is shown in

FIGURE 9.13 Illustration of the modulation theorem: (a) a transient data signal and its Fourier spectrum

magnitude; (b) amplitude-modulated signal and its Fourier spectrum magnitude; (c) a sinusoidal data signal; and (d) amplitude modulation by a sinusoidal signal.

Figure 9.14 Note that, in practice, to achieve satisfactory modulation, other components such assignal preamplifiers and filters will be needed.

There are many applications of amplitude modulation In some applications, modulation isperformed intentionally In others, modulation occurs naturally as a consequence of the physicalprocess, and the resulting signal is used to meet a practical objective Typical applications ofamplitude modulation include the following:

1 Conditioning of general signals (including DC, transient, and low-frequency) by ing the advantages of AC signal conditioning hardware

exploit-2 Improvement of the immunity of low-frequency signals to low-frequency noise

3 Transmission of general signals (DC, low-frequency, etc.) by exploiting the advantages

of AC signals

4 Transmission of low-level signals under noisy conditions

5 Transmission of several signals simultaneously through the same medium (e.g., sametelephone line, same transmission antenna, etc.)

6 Fault detection and diagnosis of rotating machinery

The role of amplitude modulation in many of these applications should be obvious if one stands the frequency-shifting property of amplitude modulation Several other types of applicationsare also feasible due to the fact that the power of the carrier signal can be increased somewhatarbitrarily, irrespective of the power level of the data (modulating) signal The six categories ofapplications mentioned above will be discussed — one by one

under-AC signal-conditioning devices such as under-AC amplifiers are known to be more “stable” than their

DC counterparts In particular, drift problems are not as severe and nonlinearity effects are lower

in AC signal-conditioning devices Hence, instead of conditioning a DC signal using DC hardware,one can first use the signal to modulate a high-frequency carrier signal Then, the resulting high-frequency modulated signal can be conditioned more effectively using AC hardware

The frequency-shifting property of amplitude modulation can be exploited in making frequency signals immune to low-frequency noise Note from Figure 9.13 that by amplitudemodulation, the low-frequency spectrum of the modulating signal can be shifted out into a very

low-high-frequency region by choosing the carrier frequency f c sufficiently large Then, any

low-frequency noise (within the band 0 to f c – f b) would not distort the spectrum of the modulated

signal Hence, this noise can be removed by a high-pass filter (with cutoff at f c – f b) without affectingthe data Finally, the original data signal can be recovered by demodulation Note that the frequency

of a noise component can very well be within the bandwidth f b of the data signal and, hence, ifamplitude modulation is not employed, noise can directly distort the data signal

Transmission of AC signals is more efficient than that of DC signals Advantages of ACtransmission include lower energy dissipation problems Hence, a modulated signal can be trans-mitted over long distances more effectively than can the original data signal alone Furthermore,transmission of low-frequency (large wavelength) signals require large antennas Hence, when

FIGURE 9.14 Representation of an amplitude modulator.

amplitude modulation is employed (with an associated reduction in signal wavelength), the size ofbroadcast antenna can be effectively reduced.

Transmission of weak signals over long distances is not desirable because further signalweakening and corruption by noise could produce disastrous results By increasing the power ofthe carrier signal to a sufficiently high level, the strength of the modulated signal can be elevated

to an adequate level for long-distance transmission

It is impossible to simultaneously transmit two or more signals in the same frequency rangeusing a single telephone line This problem can be resolved using carrier signals with significantlydifferent carrier frequencies to amplitude-modulate the data signals By picking the carrier frequen-cies sufficiently farther apart, the spectra of the modulated signals can be made non-overlapping,thereby making simultaneous transmission possible Similarly, with amplitude modulation, simul-taneous broadcasting by several radio (AM) broadcast stations in the same broadcast area hasbecome possible

Fault Detection and Diagnosis

One use of the amplitude modulation principle that is particularly important in the practice ofmechanical vibration is in the fault detection and diagnosis of rotating machinery In this method,modulation is not deliberately introduced, but rather it results from the dynamics of the machine.Flaws and faults in a rotating machine are known to produce periodic forcing signals at frequencieshigher than, and typically at an integer multiple of, the rotating speed of the machine For example,backlash in a gear pair will generate forces at the tooth-meshing frequency (equal to the product:number of teeth × gear rotating speed) Flaws in roller bearings can generate forcing signals atfrequencies proportional to the rotating speed times the number of rollers in the bearing race.Similarly, blade passing in turbines and compressors, and eccentricity and unbalance in a rotor,can produce forcing components at frequencies that are integer multiples of the rotating speed.Now, the resulting vibration response will be an amplitude modulated signal, where the rotatingresponse of the machine modulates the high frequency forcing response This can be confirmedexperimentally by Fourier analysis (fast Fourier transform or FFT) of the resulting vibration signals.For a gear box, for example, it will be noticed that, instead of getting a spectral peak at the geartooth-meshing frequency, two side bands are produced around that frequency Faults can be detected

by monitoring the evolution of these side bands Furthermore, because side bands are the result ofmodulation of a specific forcing phenomenon (e.g., gear-tooth meshing, bearing-roller hammer,turbine-blade passing, unbalance, eccentricity, misalignment, etc.), one can trace the source of aparticular fault (i.e., diagnose the fault) by studying the Fourier spectrum of the measured vibrations.Amplitude modulation is an integral part of many types of sensors In these sensors a high-frequency carrier signal (typically the AC excitation in a primary winding) is modulated by themotion Actual motion can be detected by demodulation of the output Examples of sensors thatgenerate modulated outputs are differential transformers (LVDT, RVDT), magnetic-induction prox-imity sensors, eddy current proximity sensors, AC tachometers, and strain-gage devices that use

AC bridge circuits Signal conditioning and transmission will be facilitated by amplitude modulation

in these cases However, the signal has to be demodulated at the end, for most practical purposessuch as analysis and recording

9.3.3 D EMODULATION

Demodulation (or discrimination or detection) is the process of extracting the original data signal

from a modulated signal In general, demodulation must be phase sensitive in the sense that thealgebraic sign of the data signal should be preserved and determined by the demodulation process

In full-wave demodulation, an output is generated continuously In half-wave demodulation, no

output is generated for every alternate half-period of the carrier signal