Microstrip Antennas Part 14 pptx

The results presented in this chapter point out that radiation losses need to be taken into account for accurate microstrip antenna array design at mm wave frequencies.. The first part d

Microstrip Antenna Arrays 379 Fig 27 presents S11 results of the folded antenna results for different position relative to the human body Explanation of Fig 26 is given in Table 6 From results shown in Fig 13 we can see that the folded antenna has V.S.W.R better than 2.0:1 for air spacing up to 5mm between the antennas and patient body If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by 5%

Plot colureSensor position

Red Shirt thickness 0.5mm

Blue Shirt thickness 1mm

Pink Air spacing 2mm

Green Air spacing 4mm

Sky Air spacing 1mm

Purple Air spacing 5mm

Table 6 Explanation of Fig 26

ColureSensor position

Red Shirt thickness 0.5mm

Blue Air spacing body to shirt 1mm

Pink Belt thickness 4mm

Sky Air spacing shirt to belt 1mm

Green Air spacing shirt to belt 4mm

Table 7 Explanation of Fig 27

410 420 430 440 450 460 470 480 490

-18 -16 -14 -12 -10 -8 -6

-20 -4

dB(S(1,1))=-10.447400.0MHz

m2freq=

dB(S(1,1))=-14.809430.0MHzm3

freq=

dB(S(1,1))=-9.962450.0MHz

Shirt 0.5mm

Air 1mmAir 2mm

Air 4mm

1mm air between body and shirtBelt 4mm

Fig 27 S11 results for different belt thickness

S11results in Fig 27 presents the folded antenna matching when the belt thickness has been changed from 2 to 4mm Explanation of Fig 27 is given in Table 7 S11results are better than -10dB for belt thickness ranging from to 2 to 4mm Computed S11 and S22 results was better

than-10dB for different body tissues with dielectric constant ranging from 40 to 50

Computed S11 and S22 results were better than -10dB for different shirts and belts with

dielectric constant ranging from 2 to 4

4.4 Medical applications

An application of the proposed antenna is shown in Fig 28 Three to four folded dipole

antennas may be assembled in a belt and attached to the patient stomach The cable from

each antenna is connected to a recorder The received signal is routed to a switching matrix

The signal with the highest level is selected during the medical test The antennas receive a

signal that is transmitted from various positions in the human body Folded antenna may be

also attached on the patient back in order to improve the level of the received signal from

different locations in the human body Fig 29 and Fig 30 show various antenna locations on

the back and front of the human body for different medical applications In several

applications the distance separating the transmitting and receiving antennas is less than

2D²/λ, where D is the largest dimension of the source of the radiation In these applications

the amplitude of the electromagnetic field close to the antenna may be quite powerful, but

because of rapid fall-off with distance, they do not radiate energy to infinite distances, but

instead their energies remain trapped in the region near the antenna, not drawing power

from the transmitter unless they excite a receiver in the area close to the antenna Thus, the

near-fields only transfer energy to close distances from the receivers, and when they do, the

result is felt as an extra power-draw in the transmitter The receiving and transmitting

antennas are magnetically coupled Change in current flow through one wire induces a

voltage across the ends of the other wire through electromagnetic induction The amount of

inductive coupling between two conductors is measured by their mutual inductance In

these applications we have to refer to the near field and not to the far field radiation pattern

Wearable Diversity antennas

Data recorder

Belt

Fig 28 Wearable antenna

Microstrip Antenna Arrays 381

Antenna location

Fig 29 Printed Antenna locations on the back for medical applications

Antennas Antennas

Fig 30 Printed Patch Antenna locations for medical applications

In Fig 31and 32 several microstrip antennas for medical applications at 434MHz are shown

Fig 31 Microstrip Antennas for medical applications

Fig 32 Microstrip Antennas for medical applications

Microstrip Antenna Arrays 383

In Fig 31 and in Fig 32 one can see different designs of dual polarized microstrip antennas with 10% bandwidth around 434MHz The loop antenna is with a ground plane on the antenna back The loop antenna diameter is around 50mm

5 Conclusion

A 64 microstrip antenna array with efficiency of 67.6% and a 256 microstrip antenna array with efficiency of 50.47% have been presented in this Chapter Methods to reduce losses in mm-wave microstrip antenna arrays have been described in this Chapter The results presented in this chapter point out that radiation losses need to be taken into account for accurate microstrip antenna array design at mm wave frequencies By minimizing the number of bend discontinuities the gain of the 256 microstrip antenna array has been increased by 1dB

Several applications of mm wave microstrip antenna arrays have been presented Losses in the microstrip feed network form a significant limit on the possible applications of microstrip antenna arrays in mm wave frequencies MM wave microstrip antenna arrays may be employed in communication links, seekers and detection arrays The array may consist around 256 elements to 1024 elements Design considerations of the antenna and the feed network are given in this chapter Optimization of the antenna structure and feed network allows us to design and fabricate microstrip antenna arrays with high efficiency This chapter presents wideband microstrip antennas with high efficiency for medical applications The antenna bandwidth is around 10% for VSWR better than 2:1 The antenna beam width is around 100º The antenna gain is around 2dBi The antenna S11 results for different belt thickness, shirt thickness and air spacing between the antennas and human body are given in this chapter The effect of the antenna location on the human body should

be considered in the antenna design process If the air spacing between the sensors and the human body is increased from 0mm to 5mm the antenna resonant frequency is shifted by

5%.The proposed antenna may be employed in Medicare RF systems

6 References

J.R James, P.S Hall & C Wood, (1981) Microstrip Antenna Theory and Design,1981

A Sabban & K.C Gupta, (1991) Characterization of Radiation Loss from Microstrip

Discontinuities Using a Multiport Network Modeling Approach, I.E.E.E Trans on M.T.T, Vol 39,No 4, April 1991,pp 705-712

A Sabban, (1991) PhD Thesis, Multiport Network Model for Evaluating Radiation Loss and

Coupling Among Discontinuities in Microstrip Circuits, University of Colorado at Boulder, January 1991

P.B Kathei & N.G Alexopoulos, (1985) Frequency-dependent characteristic of microstrip,

MTT-33, discontinuities in millimeter-wave integrated circuits, IEEE Trans Microwave Theory Tech, vol pp 1029-1035, Oct 1985

A Sabban, (1983) A New Wideband Stacked Microstrip Antenna, I.E.E.E Antenna and

Propagation Symposium, Houston, Texas, U.S.A, June 1983

A Sabban & E Navon (1983) A MM-Waves Microstrip Antenna Array, I.E.E.E Symposium,

Tel-Aviv, March 1983

A Sabban, (1981) Wideband Microstrip Antenna Arrays, I.E.E.E Antenna and Propagation

Symposium MELCOM, Tel-Aviv,1981

M M Milkov, (2000) Millimeter-Wave Imaging System Based on Antenna-Coupled

Bolometer, MSc Thesis, UCLA UCLA (2000)

G de Lange et al., (1999) A 3*3 mm-wave micro machined imaging array with sis mixers,

Appl Phys Lett 75 (6), pp 868-870 (1999)

A Rahman et al., (1996) Micromachined room temperature microbolometers for mm-wave

detection, Appl Phys Lett 68 (14), pp 2020-2022 (1996)

A Luukanen et al., US Patent 6242740 (2001)

M D Jack et al., (2001) US Patent 6329655 (2001)

G N Sinclair et al., (2000) Passive millimeter wave imaging in security scanning, Proc

SPIE Vol 4032, pp 40-45, (2000)

G Kompa & R Mehran, (1975) Planar waveguide model for computing microstrip

components, Electron Lett., vol 11, no 9, pp 459-460, 1975

Lawrence C Chirwa; Paul A Hammond; Scott Roy & David R S Cumming, (2003)

Electromagnetic Radiation from Ingested Sources in the Human Intestine between

150 MHz and 1.2 GHz, IEEE Transaction on Biomedical eng., VOL 50, NO 4, pp

484-492, April 2003

D.Werber; A Schwentner & E M Biebl, (2006) Investigation of RF transmission properties

of human tissues, Adv Radio Sci., 4, pp 357–360, 2006

16

Microstrip Antennas for Indoor Wireless

Dynamic Environments

Mohamed Elhefnawy and Widad Ismail

Universiti Sains Malaysia (USM),

Malaysia

1 Introduction

This chapter is organized in two parts The first part deals with the design and implementation of a microstrip antenna array with Butler matrix The planar microstrip antenna array has four beams at four different directions, circular polarization diversity, good axial ratio, high gain, and wide bandwidth by implementing the 4×4 Butler matrix as a feeding network to the 2×2 planar microstrip antenna array The circular polarization diversity is generated by rotating the linearly polarized identical elements of the planar

microstrip antenna array so that the E-field in the x-direction is equal to the E-field in the

y-direction Then, by feeding the planar array with Butler matrix, phase delay of π/ 2

between those two E-fields is provided

In the second part of this chapter, the analysis, design and implementation of an Aperture Coupled Micro-Strip Antenna (ACMSA) are introduced A quadrature hybrid is used as a feeder for providing simultaneous circular polarization diversity with a microstrip antenna; but the utilization of the quadrature hybrid as a feeder results in large antenna size In order

to minimize the antenna size, the microstrip antenna is fed by a quadrature hybrid through two orthogonal apertures whose position is determined based on a cavity model theory The size of the proposed ACMSA is small due to the use of the aperture coupled structure The cavity model theory is started with Maxwell's equations, followed by the solution of the homogeneous wave equations Finally, the eigenfunction expansion for the calculation of the input impedance is presented This chapter is organized as follows The first part deals with design and implementation of a microstrip antenna array with Butler matrix, which describe the design details of a rectangular microstrip patch antenna and a 4×4 Butler matrix Further, analysis of planar microstrip antenna array with Butler matrix and the development of the radiation pattern for the planar microstrip antenna array are presented

In the second part, the design and implementation of an aperture coupled microstrip antenna, the analysis of ACMSA using cavity model, the circular polarization diversity with ACMSA and the geometry of the ACMSA are described

2 Design and implementation of the microstrip antenna array with Butler matrix

A planar microstrip antenna array with a Butler matrix is implemented to form a microstrip antenna array that has narrow beamwidth, circular polarization and polarization diversity

This microstrip antenna array improves the system performance in indoor wireless dynamic

environments A circularly polarized microstrip antenna array is designed such that it

consists of four identical linearly polarized patches A 2×2 planar microstrip antenna array

and a 4×4 Butler matrix are designed and simulated using advanced design system and

Matlab software The measured results show that a combination of a planar microstrip

antenna array and a 4×4 Butler matrix creates four beams two of which have RHCP and the

other two have LHCP

2.1 Design of the rectangular microstrip patch antenna

A rectangular microstrip patch antenna is designed based on the Transmission Line Model

(TLM) in which the rectangular microstrip patch antenna is considered as a very wide

transmission line terminated by radiation impedance Figure 1 shows a rectangular

microstrip patch antenna of length L and width W Ms is the magnetic current of each

radiating slot of the microstrip patch antenna and s is the width of each radiating slot

Fig 1 Inset fed rectangular microstrip patch antenna

Figure 2 shows the transmission line model of the antenna where GR and CF represent the

radiation losses and fringing effects respectively The input impedance of an inset fed

rectangular microstrip patch antenna is given by the equation (1) [1]

1 cos2

o in

R

x Z

where Xo is the distance into the patch, G12 is the coupled conductance between the

radiating slots of the antenna [2]

W Feed

Microstrip Antennas for Indoor Wireless Dynamic Environments 387

Fig 2 Transmission line model of the rectangular microstrip patch antenna

The inset fed rectangular microstrip patch antenna is designed using Matlab software based

on the expression for the input impedance which is given by equation (1) The input impedance depends on the microstrip line feed position as shown in Figure 3

Fig 3 Dependence of the input impedance on the distance into the patch

G R

L< λ/2

Slot 2 Slot 1

2.2 Design of the 4×4 Butler matrix

The Butler matrix is used as a feeding network to the microstrip antenna array and it works

equally well in receive and transmit modes The 4×4 Butler matrix as shown in Figure 4

consists of 4 inputs, 4 outputs, 4 hybrids, 1 crossover to isolate the cross-lines in the planar

layout and some phase shifters [3] Each input of the 4 4× Butler matrix inputs produces a

different set of 4 orthogonal phases; each set used as an input for the four element antenna

array creates a beam with a different direction The switching between the four Butler inputs

changes the direction of the microstrip antenna array beam

Advanced design system (ADS) has been used for simulating the 4×4 Butler matrix as

shown in Figure 5 Table 1 shows a summary of the simulated and the measured phases that

are associated with the selected port of the 4×4 Butler matrix

Fig 4 4×4 Butler matrix geometry

Phase A Antenna 1 Antenna 2 Phase B Antenna 3 Phase C Antenna 4 Phase D

Microstrip Antennas for Indoor Wireless Dynamic Environments 389

1

MLIN TL5

L=27.637 mm

MLIN TL6

L=16.9601 mm

1 1

2

Term Term7

Z=50 Ohm Num=7

Step=1.0 MHz Stop=3 GHz Start=1.4 GHz

Z=50 Ohm Num=8

1 1

2

Term Term6

Z=50 Ohm Num=6

1

1

2

Term Term4

Z=50 Ohm Num=4

1

2

Term Term3

Z=50 Ohm Num=3

Z=50 Ohm Num=1

2hybrid_real X21

1

2

MLIN TL66

L=6 mm

1

2

MLIN TL67

L=6 mm

1

2

MLIN TL64

1

2

3 4

2hybrid_real X20 1

2

3 4

2cross_over_real X22

Fig 5 ADS schematic for 4×4 Butler matrix

2.2.1 Design of quadrature hybrid

The design of a quadrature hybrid is based on the scattering matrix equation which can be

obtained as follows [4]:

0 1 0

0 0 11

1 0 02

0 1 0

Hybrid

j j S

j j

The design of a quadrature hybrid using ADS is started by creating ADS schematic for the

quadrature hybrid using ideal transmission lines as shown in Figure 6 [3]

Fig 6 ADS schematic for an ideal quadrature hybrid

1

Port

P2Num=2

TLIN

TL2F=2.437 GHzE=theta_tbZ=zo_tb Ohm

12

TLIN

TL3F=2.437 GHzE=theta_lrZ=zo_lr1

2

TLIN

TL4F=2.437 GHzE=theta_lrZ=zo_lr

Microstrip Antennas for Indoor Wireless Dynamic Environments 391 Then a new ADS schematic for the quadrature hybrid is created using a microstrip substrate element (MSUB) and real transmissions lines as shown in Figure 7

Figure 8 shows the higher level ADS schematic which is used to optimize and modify the dimensions of the real quadrature hybrid to emulate the ideal quadrature hybrid as closely

as possible

The simulated S-parameters of the real quadrature hybrid indicate that the power entering

in any port is not reflected, and it is equally divided between two output ports that are existed at the other side of the hybrid, while no power is coupled to the port which is existed

at the same side of the input port This quadrature hybrid can operate over a bandwidth from 1.9 GHz to 2.8 GHz as shown in Figure 9

Fig 7 ADS schematic for a real quadrature hybrid

2

3 1

MTEE_ADS Tee3

W3=W_lr mm W2=W_tb mm W1=W_50 mm Subst="MSub1"

1 2

MLIN TL2

L=l_tb mm W=W_tb mm Subst="MSub1"

1

2

MLIN TL4

L=l_lr mm W=W_lr mm Subst="MSub1"

1

2

MLIN TL3

L=l_lr mm W=W_lr mm Subst="MSub1"

2 3 1

MTEE_ADS Tee2

W3=W_lr mm W2=W_50 mm W1=W_tb mm Subst="MSub1"

1 2

MLIN TL1

L=l_tb mm W=W_tb mm Subst="MSub1"

2

3

1

MTEE_ADS Tee1

W3=W_lr mm W2=W_tb mm W1=W_50 mm Subst="MSub1"

l_tb=14.4394 {o}

W_lr=1.67935 {o}

W_tb=2.8446 {o}

Eqn Var

MTEE_ADS Tee4

W3=W_lr mm W2=W_50 mm W1=W_tb mm Subst="MSub1"

1

Port P3

Num=3

1

Port P2

Num=2

RangeMax[1]=2.45 GHZ RangeMin[1]=2.42 GHZ RangeVar[1]="freq"

Weight=1 Max=.0001 Min=0 SimInstanceName="SP1"

Term Term8

Z=50 Ohm Num=8

1 2

Term Term7

Z=50 Ohm Num=7 1

1 2

Term Term5

Z=50 Ohm Num=5

1

1 2

Term Term6

Z=50 Ohm Num=6 1

1

1

1 2

Term Term2

Z=50 Ohm Num=2 1

2

Term Term3

Z=50 Ohm Num=3 1

S-PARAMETERS Optim

Optim1

SaveCurrentEF=no UseAllGoals=yes UseAllOptVars=yes SaveAllIterations=no SaveNominal=no UpdateDataset=yes SaveOptimVars=no SaveGoals=yes

The crossover provides an efficient mean to isolate two crossing transmission lines The

design of the crossover depends on the following scattering matrix equation [5]

j j

Microstrip Antennas for Indoor Wireless Dynamic Environments 393

-40 0

-40 0

TLIN

TL1F=2.437 GHzE=theta_tbZ=zo_tb Ohm

TLIN

TL2F=2.437 GHzE=theta_tbZ=zo_tb Ohm

TLIN

TL10F=2.437 GHzE=theta_tbZ=zo_tb Ohm

12

TLIN

TL11F=2.437 GHzE=theta_tbZ=25 Ohm

12

TLIN

TL3F=2.437 GHzE=theta_lrZ=zo_lr

Num=2

1

Port P3