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The characteristics of the output signal to a digital subsystem are:X spectrum baseband, with bandwidth up to 20 MHz X dynamic range reduced by AGC to meet requirements of the ADC In doi

Part II

Front End Technology

Front End design – including RF Architecture, Data Conversion and Digital Front Ends – hasemerged as a key issue as SDR techniques are finding themselves increasingly embodied bystealth into today’s new products

The radical solution – ‘Pure’ Software Radio, with A/D conversion at the antenna – is notyet feasible at GHz carrier frequencies However, recent technology advances suggest it may

be nearer than had been thought

Edited by Walter Tuttlebee Copyright q 2002 John Wiley & Sons, Ltd ISBNs: 0-470-84318-7 (Hardback); 0-470-84600-3 (Electronic)

This chapter is structured in four parts Initially, we gather data to define the requirementsfor the illustrative design of SDR hardware for commercial wireless applications In thesecond part, we attempt to define the problems that are associated with the design of SDRhardware, both the receiver and the transmitter aspects In the third, we consider techniqueswhich may be of value in solving these problems before finally drawing some conclusions.

In considering these requirements, the chapter is based around the proposition that ourSDR must be able to process the major European air interface standards The performancerequirements that these standards demand are then surveyed

Basic receiver design is considered by looking first at the architectural issues The pros andcons of single and multiple conversion architectures are then discussed The important issue

of circuit linearity, and its relationship with blocker specifications, is examined Alternativespecifications for radio linearity are defined, and the reason for their use is briefly explained

‘Signal budget,’ as a basis for the receiver design, is discussed An example is given to showhow performance specifications relate to this budget, and the consequent performance speci-fication of individual circuit elements Image signals and the problems they cause in the

1 The term ‘ideal SDR’ used in this chapter should not be confused with the concept of the ‘pure’ software radio The latter refers to a software radio with A/D conversion at the carrier frequency; the former refers to a fully flexible and dynamically reconfigurable ‘pragmatic’ software radio, still incorporating radio frequency translation stages prior to the A/D converter and digital processing subsystem.

Copyright q 2002 John Wiley & Sons, Ltd ISBNs: 0-470-84318-7 (Hardback); 0-470-84600-3 (Electronic)

design of a superheterodyne SDR receiver are then covered The relationship of the imagesignal to the IF frequency, as well as the relationship of the required image rejection to thereceiver blocker specifications, is also explained.

The design of an SDR transmitter is introduced by explaining that the filter problemswithin a receiver mirror their analogs within the SDR transmitter Transmitter architecturesare examined, and similar conclusions are drawn about their architectures as were drawnabout the receiver Transmitter linearity and the related efficiency issues are highlighted andtechniques of PA linearization briefly summarized The zero IF stage is examined compre-hensively, including a detailed description of the specific issues which arise from this archi-tecture Flexible filtering is then discussed and a brief summary of the ways in which it could

be realized, or has been realized, is presented Emphasis is placed on ensuring that the tuningdoes not introduce nonlinearities into the circuit performance and an introduction is presentedinto the potential role that micro-electro-mechanical system (MEMS) technology could play

in future designs

The body of the chapter concludes with an examination of the ‘Low IF’ design as a possiblecompromise between the superheterodyne and zero IF architectures

2.1 Requirements and Specifications

There are three driving forces for the development of SDR The first impetus derives from therequirement that a mobile phone can provide ‘world roaming.’ This means that the phone, aswell as being able to operate in Europe to the GSM radio standard, should be able to operate

in the United States to their IS54 and IS95 systems, and in Asia and Japan with their PDC andPHS systems The second stimulus revolves around trying to combine the performancefeatures of a radiotelephone (GSM, DECT, and UMTS), with the functionality of a personalarea network (PAN), (e.g Bluetooth), and that of a local area Network (LAN) (e.g HIPER-LAN) The third motivation is that SDR could drive down the production costs through thescale economies of a single radio platform for multiple standards and hence markets.Improvements could be made via ‘software upgrades’ and the radio could be ‘future proofed’

to some degree

In this section we review the radio specifications of some of the major European nications standards to define the performance that will be required of an SDR capable ofencompassing all these air interface standards It would be possible to spread the net widerand also embrace North American and Asian standards; however, the European standards aresufficiently representative to highlight all the major issues involved [1–8]2

commu-2.1.1 Transmitter Specifications

The most important design parameters when dealing with SDR transmitter design are:

X output power level

X power control range

X spurious emissions

2 These standards embrace standard voice telephony (GSM and DECT), 3G standards (UMTS), personal area networks (Bluetooth), and local area networks (HIPERLAN/2).

Frequency of operation is also the other obvious parameter – this is discussed following theseparate transmitter and receiver specification discussions, with requirements summarizedlater in Figure 2.2.

2.1.1.1 Transmitter Output Power Levels

The power levels to be provided by a mobile station depend on the standard and on its class

In all cases, the transmitter should provide output power control over a significant range tocomparatively fine tolerances This will impose critical challenges on the architecture used.Table 2.1 summarizes this information

2.1.1.2 Spurious Emission Specifications

All air interface standards specify spurious emission with a mask specification They arebest summarized graphically A graphical summary of all ‘in-band’ spurious emissionspecifications is included in Appendix A1 to this chapter Out-of-band emissions are alsospecified in the relevant standards For information on these the reader is referred toreferences [1–8]

2.1.2.1 Input Sensitivity and Maximum Input Level

Table 2.2 summarizes the sensitivity requirements of our target group of air interface dards

stan-2.1.2.2 Blocker Specifications

All air interface standards specify blocker signal levels with a mask type of specification.Again, this is best summarized graphically and is included as Appendix A2 to this chapter

2.1.3 Operating Frequency Bands

Table 2.3 lists the frequency bands for the air interface standards considered in this chapter;this information is also shown graphically in Figure 2.1

2.2 Receiver Design Considerations

2.2.1 Basic Considerations

The basic receiver function is to take a real, low power, RF signal and down-convert it to acomplex (in-phase and quadrature, I/Q) baseband signal During this process, the signalpower level is increased The following list describes the characteristics of the input signal

to a hypothetical SDR receiver and the output signal from that receiver

The characteristics of the input signal are:

X high dynamic range up to 2 15 dBm

X spectrum band pass, with center frequencies varying from

876 MHz to 5725 MHz

Table 2.2 Input signal level specifications

Air interface standard Reference

sensitivity level(dBm)

Maximum input level (dBm)

GSM 900 Small MS 2 102 2 15

Other MS 2 104DCS 1800 Class 1 or Class 2 2 100/ 2 102 2 23

Class 3 2 102PCS 1900 Normal 2 102 2 23

The characteristics of the output signal (to a digital subsystem) are:

X spectrum baseband, with bandwidth up to 20 MHz

X dynamic range reduced by AGC to meet requirements of the ADC

In doing this, the receiver must

X keep the signal power sufficiently greater than the noise power, to ensure the outputsignal-to-noise ratio is sufficiently high to allow appropriate BER performance of themodulation scheme used;

X ensure that high power input signals do not overload components of the receiver;

X ensure that high power nearby signals (blockers) do not effect detection of the wantedsignal;

X ensure that signals of the wanted frequency can be separated from signals at the image3frequency

The first two points on this list are generally accommodated by careful design The latter twopoints are problems that can be tackled by selection of an appropriate architecture and

3

Image signals are discussed further under ‘Image Rejection’ within this section on receiver design.

Table 2.3 Frequency of operation of major European air interface standards

Air interface standard Uplink (MHz) Downlink (MHz) Duplex spacing

(MHz)GSM 900 890–915 935–960 45

most other countries

2400–2483.5 2400–2483.5 –Spain 2455–2475 2455–2475 –

France 2446.5–2435 2446.5–2435 –

HIPERLAN\2 5150–5350 5150–5350

5470–5725 5470–5725 –

application of appropriate technological ‘fixes’ such as image reject mixing, linearization,and variable preselect filters.

Important commercial requirements, which place constraints on this, are:

X ability to be manufactured as an integrated circuit, with a minimum of external nents;

compo-X low power consumption to allow portable operation with long battery life

The next section discusses the comparative advantage of various receiver architectures;subsequent sections describe important considerations in practical receiver design

Figure 2.1 Diagrammatic representation of the operating frequencies of the major European airinterface standards (excluding HIPERLAN/2)

2.2.2 Receiver Architectures

The primary distinction between receivers is the number of stages taken to down-convert asignal to baseband Direct conversion takes one down-conversion; superheterodyne receiversemploy two or more In general, complexity increases with the number of down-conversions

As we explore alternative architectures it will be shown that the simplicity of direct sion brings with it several technical problems which would appear to make direct conversionarchitecture inappropriate for an SDR receiver These issues are treated in more detail later inthe chapter

conver-2.2.2.1 Direct Conversion Architecture

A basic direct conversion receiver architecture is shown in Figure 2.2 This receiver consists

of a low noise amplifier (LNA) which provides modest RF gain at a low noise figure Theoutput signal from the LNA is filtered in a preselect filter, and down-converted in a complex(I,Q) mixer The majority of the gain and automatic gain control (AGC) is provided in a highgain baseband amplifier

Its advantages are:

X low complexity

X suitable for integrated circuit realization

X simple filtering requirements

X image signal suppression is easier (compared to multiple conversion architecture)Its disadvantages are:

X A local oscillator is required, in which the two output signals are accurately in phasequadrature and amplitude balance, over a frequency range equal to the frequency range

of the input signal

Figure 2.2 Direct conversion receiver architecture

X The mixers needs to be balanced and to be able to operate over a correspondingly widefrequency band.

X Local oscillator leakage through the mixer and LNA will be radiated from the antennaand reflected back into the receiver from that antenna The reflected signal will vary withthe physical environment in which the antenna is placed This ‘time varying’ DC offsetcaused by ‘self-mixing’ is a problem

X Most of the signal gain occurs in one frequency band creating the potential for ity

instabil-X 1/f noise is a major problem

X Second order distortion product mix down ‘in-band’

All of these points are explained in more detail later in the chapter

2.2.2.2 Multiple Conversion Architecture

A multiple conversion receiver is shown in Figure 2.3

Its advantages are:

X good selectivity (due to the presence of preselect and channel filters;

X gain is distributed over several amplifiers operating in different frequency bands;

X conversion from a real to a complex signal is done at one fixed frequency; therefore aphase quadrature, amplitude balanced, local oscillator is only required at a singlefrequency

Its disadvantages are:

X the complexity is high;

X several local oscillator signals may be required;

X specialized IF filters are required; this makes it impossible to achieve single chip zation of a superheterodyne receiver

reali-Figure 2.3 Multiple conversion superheterodyne architecture

Although the multiple conversion stage of Figure 2.3 only shows two explicit conversions (one in the RF hardware and one in digital signal processing (DSP)), furtherconversions can be done in the DSP via the processes of ‘decimation’ and/or ‘sub-sampling’ Such a receiver architecture may represent the best choice for an SDR receiverdesign today, given that the two principal disadvantages4of direct conversion are practicallyinsurmountable for a wideband SDR application with current technology With this archi-tecture, the first conversion may be done in RF hardware, and all of the others are done inDSP.

down-2.2.2.3 Low IF Architecture

Use of the frequency conversion architecture with a low IF represents a pragmatic attempt tocombine the advantages of a superheterodyne structure with the advantages of a directconversion architecture (see [9,10]) Having a low IF means that the image rejection require-ments are not as onerous as with the superheterodyne structure, and the fact that the LO signal

is not the same frequency as the wanted signal minimizes the DC offset problems inherent inthe direct conversion architecture

Its advantages are:

X the DC Offset problems associated with direct conversion architecture can be overcomewhile retaining most of the benefits of this architecture;

X lower complexity than the superheterodyne approach (but slightly greater than the directconversion)

Its disadvantages are:

X better image rejection is required from a low IF receiver than that required of the directconversion receiver

2.2.3 Dynamic Range Issues and Calculation

In this section we develop equations that are essential for the design of receivers and mitters for SDR applications

trans-2.2.3.1 Third-order Distortion Components and Third-Order Intercept

Figure 2.4 shows the relationship of the output power of the fundamental signal component,and the third-order distortion component, of an RF device, as the input signal power level isincreased This plot is typical for most nonlinear devices that make up a receiver or trans-mitter (although, of course, the power levels will be different) Two features may be observedfrom this characteristic

The first feature is that the third-order distortion component increases at three times the rate

at which the fundamental component increases This is because the power in the third-ordercomponent is proportional to the cube of the input power

4

Local oscillator balance and DC offset.

Second, were it not for saturation at the output, the third-order distortion product wouldeventually reach the same power as the fundamental component This point is known asthe ‘third-order intercept’ (TOI).5The TOI of the example of Figure 2.4 can be seen to be

144 dBm

A typical spectrum analyzer display showing the results of a two-tone test6is shown inFigure 2.5 The third-order components that will appear ‘in-band’7are those at frequencies2f12 f2 and 2f22 f1 These components increase at three times the rate of the fundamentalcomponents

Figure 2.4 Illustration of the concept of third-order intercept

5 An alternative terminology for third-order intercept is IP 3 (intercept point – third-order) This is a useful notation when dealing with intercept points of other orders (second-order intercept point IP 2 ).

6 The ‘two tones’ refers to the two input signals, at frequencies f 1 and f 2

7 Other components such as 2f 1 1 f 2 will appear a long way out-of-band, and will thus not pass through the IF filtering.

With reference to Figure 2.5 again, it can be shown that:

TOI dBmð Þ ¼ P1oðdBmÞ 1 AðdBÞ

Equation (1) gives us a convenient way of calculating the TOI given a spectrum analyzerdisplay of the results of a two-tone test The measurements required are the power of the two-tone test signals, Po(dBm), and the difference between the power of the two test signals, andthe third-order distortion components, A (dB)

2.2.3.2 Cascading Devices with Known TOI and Noise Figure

The noise performance of the receiver will never be better than the input stage noise figureand the distortion performance will never be better than the final stage output TOI.When designing a receiver chain it can be difficult to know where to put effort intoimproving TOI and noise figure For example, although it is obvious that the noise figure

of the first stage is crucial, by how much can we allow the noise figures of subsequent stages

to deteriorate, without greatly degrading the overall noise performance of the receiver? Bythe same token, although we realize that the distortion performance of the final amplifier iscrucial in setting the overall distortion performance of the receiver, what distortion perfor-mance is required of the earlier stages in the chain?

A cascade connection of amplifiers is shown in Figure 2.6 The overall noise figure of such

a cascade of amplifiers is given by

1TOI2G4G3 1

1TOI3G4 1

1TOI4 1 …

of the overall distortion performance of an amplifier chain; see [12], pp 219–232 and pp.367–371 Although Equations (2) and (3) have been developed for amplifiers, they areequally applicable to lossy elements such as filters and mixers.

2.2.3.3 Relationship of Distortion Performance of an SDR Receiver Chain to BlockerSpecifications

Dynamic range describes the ability of a receiver to receive a small wanted signal in thepresence of a nearby (in frequency), large, unwanted signal One of the unique problems of anSDR receiver is the necessity of keeping the input RF bandwidth as wide as possible, toaccommodate wide bandwidth signals (e.g universal mobile telecommunications service(UMTS) with a bandwidth of 5 MHz), while at the same time preventing high power inter-fering signals affecting narrow band channels (e.g GSM style radios with a bandwidth of 200kHz) As it is practically impossible to introduce a very narrow bandwidth filter at RFfrequencies, then the linearity of the RF stage must be maintained until a filter stage isreached that will allow a narrow band channel to be selected from the wideband channel.The worst case situation arises when a narrow band channel, such as GSM, is exposed toblockers over the wideband UMTS channel A graphical interpretation of the blockingspecifications for a GSM channel is shown in Figure 2.7

A situation that will lead to blockers having an effect on the wanted channel is illustrated inFigure 2.8 Here, two high power blockers are assumed to be present in channels separatedfrom each other and from the wanted channel by 2.4 MHz8 A third-order product will beproduced within the wanted channel due to the blockers being distorted by the implicitnonlinearity of the receiver hardware It can be seen from Figure 2.7 that the blockers arepermitted to have an amplitude of up to 2 23 dBm

8 2.4 MHz is chosen because it is the maximum separation of blockers from the wanted signal, and from each other, that can fit inside the 5 MHz UMTS bandwidth (allowing for 200 kHz of the wanted signal) This number is not critical, as the blockers need to be well outside the GSM frequency band before their allowed power jumps to 0 dBm.

Figure 2.6 Cascade connection of amplifiers

The cochannel interference specification for a GSM system demands that the carrier tointerference (C/I) ratio is at least 9 dB This implies that, with an input signal of 3 dB abovethe reference level ( 2 101 dBm in the case of a GSM mobile station), the cochannelinterference is required to be less than 2 110 dBm.

If the blockers are at the maximum level of 2 23 dBm, then the difference between the twotones at a power level of 2 23 dBm and the distortion products (at 2 110 dBm), is 87 dB.This figure can be now substituted into Equation (1) to derive the required input TOI of thereceiver as

Figure 2.7 GSM blocker specifications

Figure 2.8 Scenario of blockers producing in-band third-order products

TOIin ¼ 223 1 87

The output TOI is calculated by adding the receiver gain (in dB) to the input TOI tions (in dBm) The further we progress down the receiver chain before inserting a filter toremove these blockers, the greater the required TOI of the receiver As it stands, an input TOI

specifica-of 20.5 dBm is a very demanding linearity requirement

The example shown in Figure 2.9 illustrates the above point As before, the blockers areassumed to be at a level of 2 23 dBm and the wanted signal at a level of 2 101 dBm Weassume the low noise amplifier (LNA) has a gain of 20 dB With reference to the previouscalculation, it can be seen that an input TOI of 1 20.5 dBm or an output TOI of 1 40.5 dBmwill maintain the 9 dB C/I ratio The wanted signal will be amplified to a level of 2 81 dBm,and the blockers will generate an ‘in-band’ distortion component of 2 90 dBm This leaves aSINAD figure at the output of both LNAs of 9 dB

We now consider the effect of inserting a narrowband filter in the right-hand path of Figure2.9 It is assumed that this filter has a sufficiently sharp cut off to prevent the blocking signalsfrom reaching the nonlinearity of the IF amplifier No new distortion products are producedand the SINAD at the output of the right-hand signal path is preserved at 9 dB The distortionoutput of the left-hand signal path is a combination of the distortion product of the LNAamplifier by the IF amplifier, plus the distortion product of the IF amplifier itself This effecthas been shown to be equivalent to an effective TOI (TOIout) (see Equation (3)) given by

1TOILNAGIF 1

1TOIIF

where TOILNAis the TOI of the LNA (W), TOIIFis the TOI of the IF amplifier (W) and GIFisthe power gain of the LNA (linear ratio) This yields a distortion component of 1 11 dBmand a SINAD of 2 50 dB, which would be unworkable Note the TOI of the LNA is biggerthan it needs to be for the signal path on the left-hand side of Figure 2.9 9; however, it isneeded to achieve reasonable distortion performance for the example on the right-hand side(even when a channelization filter is included).

2.2.3.4 Measurement and Calculation of Dynamic Range: Spurious Free DynamicRange (SFDR)

Maximizing the dynamic range of a receiver implies that the receiver has a maximum TOIand a minimum noise figure Improving one of these parameters, at the expense of the other,will not increase the dynamic range of the receiver One measure of the dynamic range is thespurious free dynamic range (SFDR) The SFDR measures the difference in power betweenthe noise floor, and a signal power that would just cause a third-order distortion component toemerge from the noise

It can be shown that the SFDR is given by

Examination of Equation (5) shows that SFDR is determined by the difference between theTOI and the noise figure (NF) So a large SFDR is obtained by having a low NF and a highTOI Equation (5) can be rewritten to give the SFDR at the input as

2.2.4 Adjacent Channel Power Ratio (ACPR) and Noise Power Ratio (NPR)

The broadband nature of the signals used in modern radio systems, combined with the closespacing of the channels, has produced important changes in the way of characterizing distor-tion The TOI figure of merit is often replaced, or at least augmented, by parameters thatemploy measurement techniques more directly related to the system that the engineer istrying to characterize

The adjacent channel power ratio (ACPR) is one such parameter This parameter measuresthe effect of a signal from one channel appearing in the adjacent channel ACPR is the ratio ofthe average power in the adjacent channel to the average power in the desired channel Figure2.10 shows how measurement of ACPR is calculated This is conveniently done using aspectrum analyzer PDCand PACare measured by integrating the respective desired channeland adjacent channel powers over the channel bandwidth

9 In other words, even if the TOI of the LNA were 20 dBm, the overall TOI OUT would still be about 50 dBm (actually 49.6 dBm).

Noise power ratio (NPR) is an alternative way of characterizing distortion A white noisesignal with a notch in its normally flat spectrum is applied to the system under test Distortionproducts generated by the system will tend to fill the spectral notch (see Figure 2.10) Theratio of the average power in the signal to average power in the notch is the NPR The smallerthe NPR, the greater the distortion.

TOI, ACPR, and NPR are not vastly different in their interpretation of a distortion tion For example, a system with high third-order components on a two-tone test will exhibithigh ACPR and a low NPR Note that ACPR and NPR are parameters whose value changeswith input power level, whereas TOI is a fixed single value for a particular system Thismakes TOI an invaluable tool for design calculations

situa-2.2.5 Receiver Signal Budget

With the design of any type of receiver, a good insight into the operation of that receiver can

be obtained by plotting a diagram showing the minimum and maximum signal levels as thesignal progresses down the receiver chain towards the ADC The signal level will increase as

it passes through amplifiers and active mixers, and decrease as it passes through passivefilters, mixers, and duplexers

The minimum input signal level will be specified for a particular air interface standard as a

‘reference sensitivity level’ The maximum signal level will also need to be specified TheAGC will be required to act on the high level signal, to reduce it sufficiently to fit within theADC headroom The minimum signal level will need to be maintained sufficiently above thenoise floor to achieve the required BER performance This distance will vary with the type ofmodulation used

This type of diagram can also incorporate the rise in the noise floor through the receiver.Finite noise figures mean that there is an ‘excess’ rise in the noise floor, over and above andabove what would occur through amplification

Such a signal budget diagram for a hypothetical receiver is shown in Figure 2.11 (Forclarity, the maximum input signal and its associated automatic gain control (AGC) charac-teristics have been omitted from the diagram.)

Figure 2.10 Different ways of quantifying the IMD distortion for wideband modulated or channel signals

multi-A 100 kHz signal bandwidth has been assumed for this receiver, thus the effective thermalnoise input is 2174 1 10logð105Þ dBm The signal level increases by an amount equal tothe gain, or decreases by an amount equal to the loss, as the signal progresses down thereceiver chain The noise level increases or decreases by an amount equal to the gain or lossplus an amount given by the progressive noise figure (see Equation (2)) The differencebetween the input signal-to-noise ratio and the output signal-to-noise ratio gives the overallreceiver noise figure (4.62 dB in this example).

An ADC will have a noise floor set by the quantization noise of the converter itself.Quantization noise occurs because the converter output only ever approximates the analogsignal that it is converting It can be shown that the signal-to-noise ratio of an ADC is givenby

where b is the resolution of the ADC in bits, FS is the sampling frequency and BC is thebandwidth of the channel being sampled.

We will now redraw Figure 2.11 to concentrate on issues associated with the ADC and theAGC (see Figure 2.12) No attempt has been made to show losses that occur in mixers, filtersand duplexers Focusing on the analog to digital conversion (ADC), and with reference to thisdiagram, the following points can be made

† The maximum gain of the receiver is set by the difference between the minimum inputsignal power, and the power of a signal placed at a position suitably above the noise floor

of the ADC to maintain the BER of the modulation being used The noise floor of the ADC

is set by the resolution of the ADC, combined with the maximum input signal of the ADC

† The required AGC range is set by the difference between maximum input signal to theradio and the minimum input signal to the radio (Pin(max)2 Pin(min)) dB, less the differencebetween the maximum input power to the ADC and the noise floor of the ADC (PADCmax2

nADC) dB, plus the Eb/N0for the modulation scheme used, i.e AGC range ¼ [(Pin(max)2

Pin(min))2 (PADC(min)2 nADC) 1 Eb/N0] dB

† The noise floor at the output of the receiver can be determined by either the noise floor ofthe ADC or the thermal noise floor For narrow band systems it tends to be the noise floor

of the ADC which determines the receiver noise floor For wide band systems, it is thethermal noise floor (in which case the ADC has more resolution than required)

2.2.5.1 An Approach to Receiver Design

To make a start on the design of the receiver, we need to know certain parameters Fromknowledge of these parameters, the design calculations (also listed) can be made

1 The maximum signal input level to the ADC This establishes point A in Figure 2.12

2 The maximum blocker level likely to be present in the signal at the ADC expressed in dBrelative to the minimum signal level (usually the reference signal level plus 3 dB) Thisestablishes the separation of point A from point B and hence establishes point B in Figure2.12 The required net gain of the receiver channel can now be calculated

3 The minimum signal-to-noise ratio for the narrowest band air interface standard beingused This will establish the separation of B and C in Figure 2.12 and hence establish point

C The required resolution of the ADC can now be calculated

4 Given the receiver gain calculated with the minimum input signal (from point (2)), theposition of point D can now be derived by adding this receiver gain (in dB) to themaximum input signal level (in dBm) The required AGC range can be calculated asthe difference between point D and point A

5 Given the maximum receiver gain, the position of the noise floor for a noiseless amplifiercan be calculated to give point E

6 By setting the thermal noise floor, below or equal to the quantization noise of the ADC,point F can be established Subtracting the value of point E (in dBm) from the value ofpoint F(in dBm) will give the maximum noise figure for the receiver chain (in dB).This procedure will allow for partial design of the receiver chain

2.2.5.2 Approach when using WCDMA signals

The approach to receiver design, discussed in the previous section, is modified slightly in thecase where the signal being processed is a WCDMA signal In this case, the signal is buried inthe noise and is only retrieved when the signal is despread This will not occur in the RFchain, so the function of the RF hardware is to keep the added noise low enough, so that whenthe signal is in fact despread, the signal to noise will remain high enough so that a low errorrate results The notion of a processing gain reflects the fact that the receiver can construct areceived signal from a number of samples and thus averaging the signal is possible Proces-sing gain is expressed as:

Processing gain ¼ 10:log Chipping rate

data rate

The fact that the signal is essentially buried in noise, makes a slight variation to ourgraphical representation of the receiver signal budget of Figure 2.12 Figure 2.13 showsthis variation Note that because of the wideband nature of the signal, the thermal noisefloor will almost certainly exceed the noise floor of the ADC

will be mixed down to the set IF frequency Image signals differ in frequency from the wantedsignal by twice the IF frequency.10

Image signals are removed in a conventional receiver via the use of preselect filters Thereceiver in a GSM mobile station, for instance, should be set to receive signals in the range of935–960 MHz The preselect filter could thus have a bandwidth of 25 MHz based on a centerfrequency of 947.5 MHz ((935 1 960)/2) Assuming an IF frequency of 80 MHz, the imagefrequency bands would be 1095–120 MHz (for high side mixing) and 775–800 MHz (for lowside mixing) Both of these bands would be eliminated by any preselect filter of modestperformance

Such a comparatively narrow band image filter is not a simple option with a proper SDRreceiver as the frequency band of signals it will be set to receive should be softwaredetermined, although switchable or electronically tuneable preselect filters are one possiblesolution

Figure 2.13 Received signal levels for W-CDMA

10 Twice the IF frequency higher than the IF frequency, in the case of high side mixing, and twice the IF frequency lower than the IF frequency, in the case of low side mixing.

Image reject mixing is another way of dealing with image signals It is typically used withlow IF, or zero IF, receivers.11To get an image reject mixer to function satisfactorily, the twolocal oscillator signals need to be in precise phase quadrature, and have precise amplitudebalance Zero IF and low IF receivers can get away with relatively poor image rejectionperformance Image rejection performance of the order of 40 dB is satisfactory for a directconversion receiver (where the image signal is in fact a frequency reversed version of thewanted signal) The image rejection performance required of the conventional superheter-odyne stage is, however, much greater, and will be determined by the blocker specifications.Consider the previous example with a signal at 1095 MHz, 160 MHz away from the wantedsignal (935 MHz); GSM blocker specifications allow a signal level of 0 dBm (see Figure 2.7)

at that frequency A blocking signal at this level is not allowed to introduce a signal of morethan the reference sensitivity 1 3 dB minus 9 dB (i.e 2 110 dBm) This would mean that therequired image rejection is 0 2 ( 2 110 dB) or 110 dB This degree of performance isimpractical Low IF receivers require an image rejection performance somewhere betweenthe two extremes because the image signals will result from blocker signals that are closer tothe wanted signal A further discussion of image rejection techniques is pursued in Sections2.4.2 and 2.4.3 (see also [13,14])

As the image signal differs from the wanted signal by twice the IF frequency, if the first IF

is made high enough, then the image signals will be out-of-band, even for an SDR Thus,maximizing the first IF may be a desirable option There is a limit to how high the first IFfrequency can be made however

First, it is necessary to apply some filtering to the first IF stage to at least help with thelinearity demands of the receiver and make some inroads into rejecting unwanted signals Athigh IF frequencies, the design of such filters becomes more problematic At 500 MHz, forexample, there is a choice of a surface acoustic wave (SAW) filter or a helical filter; theformer is lossy and the latter is bulky

Second, if there has only been a small conversion in the first IF, then the second IF willrequire a bigger down-conversion This will cause the image signal to ‘close in’ on the wanted

Figure 2.14 Image problems arising from a large second conversion

signal For example Figure 2.14 shows an E-GSM signal being down-converted to a 500 MHzfirst IF The local oscillator is set to perform a high side conversion and its frequency will varyfrom 1425 to 1460 MHz accordingly The resultant image of the first down-conversionappears at a frequency of 1925–1960 MHz This signal will be easily eliminated by any

RF preselect filter The second down-conversion to 20 MHz will cause problems For thisconversion, the image signal will be at 540 MHz If 80 dB image rejection is required, thenthe first IF filter must be 80 dB down at 540 Hz This places quite a restriction on the first IFfilter These specifications can be met with an SAW filter However, the effect is describedbecause it is all too easy to assume that because the image signals have been removed fromthe first down-conversion, then subsequent down-conversions will not produce imageproblems The effect could be summarized by saying that the filter prior to any down-conversion acts to provide an RF preselection function to remove image signals and thereforethis filter should be designed accordingly

2.2.7 Filter Functions within the Receiver

To summarize the points raised in this section, in any superheterodyne receiver architecture,filters are required to perform three functions

X First, they band limit the signal to the frequency of interest This function is often referred

to as ‘channelization’ and is achieved, for preference, in the baseband of the receiver

X Second, filters are used to allow the image signal to be separated from the wanted signal.This function is performed at the first opportunity in the receiver chain

X Third, filters should prevent nearby but out-of-band ‘blocker’ signals generating cient ‘in-band’ power to interfere with the wanted signal It should be noted that if thereceiver amplifier were perfectly linear, then it would not be possible for out-of-bandsignals to generate in-band products, and a filter to achieve this function would not berequired In practice, some nonlinearity exists in all amplifiers and mixers that make upthe receiver chain This means that some degree of channelization needs to occur at afairly early stage in the amplifier-mixer chain

suffi-2.3 Transmitter Design Considerations

The design of the transmitter is somewhat similar to the design of the receiver in that there areelements in the receiver design which appear, slightly disguised in format, within the design

of the transmitter We first discuss these features before moving on to issues more particularlyrelated to transmitter design

2.3.1 Filtering Analogies between Receiver and Transmitter

The three functions performed by filters in the receiver are also performed by filters in thetransmitter With the transmitter, a filter is required to define the channel, as in the receiver Afilter is required to remove the unwanted outputs of the final up-conversion; this is compar-able to removing the image signal in the receiver A filter is also required to prevent spurious

or out-of-band emissions, analogous to preventing blocking signals generating ‘in-band’interference in the receiver case Filters required to perform these various functions areplaced at corresponding points in the receiver and transmitter chains (see Figure 2.15)

High linearity is required from the high power amplifier (HPA) to prevent spurious sions, and from the low noise amplifier (LNA) to prevent blocker signals generating in-bandinterference Overall linearity of the IF and HPA for the transmitter, and the LNA and the IFamplifier for the receiver, is required to preserve the integrity of any linear modulation beingused The linearity requirement, in this regard, is unlikely to be as critical as the linearityrequirements for preventing spurious emissions and dealing with blockers It does, however,mean that the amplifiers in the IF of both the transmitter and receiver still need to be ‘nearlinear.’

emis-2.3.2 Transmitter Architectures

Basically the same choice applies to transmitter architectures as applies to receiver tures The advantages and disadvantages associated with receiver architectures more or lesstranslate to transmitters There is no advantage in having the equivalent of a low IF receiver

architec-In the transmitter this will cause the wanted sideband to be closer to the unwanted sideband,making it difficult to remove by filtering

Figure 2.15 Duality of filters functions within a superheterodyne receiver and transmitter

2.3.2.1 Direct Conversion Transmitter

A direct conversion transmitter is shown in Figure 2.16 Its advantages are

X low complexity

X suitable for integrated circuit realization

X simple filtering requirements

X image or unwanted sideband problems are more easily dealt with, than with otherarchitectures

Its disadvantages are:

X a local oscillator that is accurately in phase quadrature, an amplitude balance over a widefrequency range is required;

X final mixers have to be wide band;

X power amplifier linearization circuits will need to operate over a wide frequency band;

X local oscillator leakage through the mixer will be radiated from the antenna

2.3.2.2 Multiple Conversion

A multiple conversion architecture is shown in Figure 2.17 Its advantages are:

X Conversion from a real to a complex signal is done at one fixed frequency, and therefore

a phase quadrature, amplitude balanced, local oscillator is only required at a singlefrequency (or it may also be done in the digital signal processor (DSP))

Its disadvantages are:

X the complexity is high;

X several local oscillator signals may be required;

X specialized IF filters will be required This makes it impossible to achieve single chiprealization of a multiple conversion transmitter

Figure 2.16 Direct up-conversion conversion transmitter

Although it appears that there are only two conversions taking place in the schematic ofFigure 2.17, other conversions could be achieved in the DSP via the use of ‘digital inter-polation’.

Given the current state of technological advancement, despite its disadvantages, the heterodyne transmitter architecture has clear benefits over competing architectures at thepresent time as a basis for a SDR transmitter design

super-2.3.3 Transmitter Efficiency and Linearity

Power efficiency is a factor that is not usually taken into account when designing receivers It

is, however, a critical element in transmitter stage design for a mobile terminal, due to theimportance of maximizing the useful ‘talk time,’ given constraints of limited battery capacity.The radio transmitter, and particularly the power amplifier, has conventionally accounted for

a large share of the power consumption of a mobile terminal This may not necessarily remainthe case with a SDR, which will face additional power demands from the software side of theradio and from the ‘linear’ receiver

Power efficiency and linearity are generally conflicting requirements The power amplifierconfigurations that provide a better efficiency (class C, for instance) are those that work in astrongly nonlinear regime, giving rise to distortion On the other hand, the amplifier classesthat better fit the linearity requirements (e.g class A) do not take full advantage of the powercapabilities of the active device, and consequently have a very poor efficiency

The efficiency figure in wireless transmitter amplifiers in current use varies between 40%for the linear classes and 60% [15] for the low consumption circuits, although some highervalues have also been reported [16] If we take into account the different standards that wewould like our SDR terminal to accommodate, some of which employ constant envelopemodulations like GMSK, or time-varying envelopes like QPSK, some trade-off is requiredbetween the efficiency and linearity performance

Figure 2.17 Multiple conversion transmitter