Micowave and Millimeter Wave Technologies Modern UWB antennas and equipment Part 12 ppt

First, transmission characteristics of the on-chip meander CPW resonator fabricated using TSMC 0.18 m CMOS technology are investigated experimentally and an equivalent circuit is develo

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Port 3

r = 0.86 mm else r = 0.83 mm

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Port 3

r = 0.86 mm else r = 0.83 mm

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The calculated results are shown in Figs 13 and 14 If the criterion of reflection is -20 dB,

then the bandwidth of |S11| in the low-frequency region is rather narrow However, |S21|

is rather small and almost all of the power from port 1 is led to port 3 at high frequencies

Port isolation between ports 2 and 3 is larger than 20 dB at frequencies between 15.3 and

16.3 GHz, as shown in Fig 15

7 Confirmation of mode conversion

Mode conversion of the electromagnetic waves may occur after passing through the bend

from port 3 to port 1 because dielectric arrays are absent in the straight waveguide portion

Only the TE20 mode needs to be considered, because the TE30 mode is under the cutoff

condition below 19.6 GHz The power ratio of the TE20 to TE10 electromagnetic wave is

obtained at port 1 The calculated results are shown in Fig 16 Since the power of the TE20

mode is very low at frequencies higher than 13.1 GHz, mode conversion will not occur

without dielectric rods in the straight portion

Fig 16 Power ratio of TE20 to TE10 at port 1

8 Simple fabrication method

As shown in the previous section, holes with diameters slightly larger than the rods will be

fabricated at the top of the waveguide and the dielectric rods will be inserted (Type B, Fig

11)

Firstly, the thick Teflon rod needs to be replaced by a thin LaAlO3 rod Fig 17 shows an improved structure over that shown in Fig 12 The coordinates and radii of the dielectric rods are shown in Table 1 A thick dielectric rod will be replaced by two thin LaAlO3 rods having radii of 0.36 mm at separated by 7.9 mm The S-parameters calculated by HFSS are shown by the solid lines in Figs 18 and 19 Secondly, S-parameters are calculated for type B

in Fig 11 with two thin LaAlO3 rods inserted from the top of the waveguide The diameter for inserting three thin rods is assumed to be 0.8 mm The S-parameters calculated by HFSS are shown by the dotted lines in Figs 18 and 19 The results for the solid and dotted lines are almost the same Port isolation between ports 2 and 3 is shown in Fig 20

1mm

Port 2 Port 1

9

6 7 8

Fig 17 Improved structure of the frequency multiplexer/demultiplexer Two thin LaAlO3

rods are used to reduce reflections

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A Dual-Frequency Metallic Waveguide System 325

The calculated results are shown in Figs 13 and 14 If the criterion of reflection is -20 dB,

then the bandwidth of |S11| in the low-frequency region is rather narrow However, |S21|

is rather small and almost all of the power from port 1 is led to port 3 at high frequencies

Port isolation between ports 2 and 3 is larger than 20 dB at frequencies between 15.3 and

16.3 GHz, as shown in Fig 15

7 Confirmation of mode conversion

Mode conversion of the electromagnetic waves may occur after passing through the bend

from port 3 to port 1 because dielectric arrays are absent in the straight waveguide portion

Only the TE20 mode needs to be considered, because the TE30 mode is under the cutoff

condition below 19.6 GHz The power ratio of the TE20 to TE10 electromagnetic wave is

obtained at port 1 The calculated results are shown in Fig 16 Since the power of the TE20

mode is very low at frequencies higher than 13.1 GHz, mode conversion will not occur

without dielectric rods in the straight portion

Fig 16 Power ratio of TE20 to TE10 at port 1

8 Simple fabrication method

As shown in the previous section, holes with diameters slightly larger than the rods will be

fabricated at the top of the waveguide and the dielectric rods will be inserted (Type B, Fig

11)

Firstly, the thick Teflon rod needs to be replaced by a thin LaAlO3 rod Fig 17 shows an improved structure over that shown in Fig 12 The coordinates and radii of the dielectric rods are shown in Table 1 A thick dielectric rod will be replaced by two thin LaAlO3 rods having radii of 0.36 mm at separated by 7.9 mm The S-parameters calculated by HFSS are shown by the solid lines in Figs 18 and 19 Secondly, S-parameters are calculated for type B

in Fig 11 with two thin LaAlO3 rods inserted from the top of the waveguide The diameter for inserting three thin rods is assumed to be 0.8 mm The S-parameters calculated by HFSS are shown by the dotted lines in Figs 18 and 19 The results for the solid and dotted lines are almost the same Port isolation between ports 2 and 3 is shown in Fig 20

1mm

Port 2 Port 1

9

6 7 8

Fig 17 Improved structure of the frequency multiplexer/demultiplexer Two thin LaAlO3

rods are used to reduce reflections

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8 10 12 14 16 -40

Fig 18 Reflection coefficient |S11| (=|S22|) for low frequencies and |S11| and |S33| for high

frequencies Solid and dotted lines denote the cases for type A and type B illustrated in Fig.9,

respectively

-20 -15 -10 -5 0

Fig 19 |S21| (=|S12|) for low frequencies and |S21| and |S31| for high frequencies Solid

and dotted lines denote the cases for type A and type B illustrated in Fig.9, respectively

0 10 20 30 40

38.5 mm at frequencies below 17 GHz, so that dielectric rods are not required in the straight portion

Secondly, a sample structure for a frequency multiplexer/demultiplexer is proposed for introducing electromagnetic waves from a coaxial cable Reflection of electromagnetic wave occurs without dielectric rods in the straight portion; therefore, another rod, made of LaAlO3

or Teflon, is introduced to reduce reflection and the calculated S-parameters The bandwidths for reflections smaller than -20 dB are still narrow; however, optimization of the design may enable the bandwidth to be expanded

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Fig 18 Reflection coefficient |S11| (=|S22|) for low frequencies and |S11| and |S33| for high

frequencies Solid and dotted lines denote the cases for type A and type B illustrated in Fig.9,

respectively

-20 -15 -10 -5 0

Fig 19 |S21| (=|S12|) for low frequencies and |S21| and |S31| for high frequencies Solid

and dotted lines denote the cases for type A and type B illustrated in Fig.9, respectively

0 10 20 30 40

38.5 mm at frequencies below 17 GHz, so that dielectric rods are not required in the straight portion

Secondly, a sample structure for a frequency multiplexer/demultiplexer is proposed for introducing electromagnetic waves from a coaxial cable Reflection of electromagnetic wave occurs without dielectric rods in the straight portion; therefore, another rod, made of LaAlO3

or Teflon, is introduced to reduce reflection and the calculated S-parameters The bandwidths for reflections smaller than -20 dB are still narrow; however, optimization of the design may enable the bandwidth to be expanded

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10 References

Ansoft Corporation (2005) Introduction to the Ansoft Macro Language, HFSS v10

Benisty, H (1996) Modal analysis of optical guides with two-dimensional photonic

band-gap boundaries, Journal of Applied Physics, 79, 10, (1996) pp.7483-7492, ISSN

0021-8979

Boroditsky, M.; Coccioli, R & Yablonovitch, E (1998) Analysis of photonic crystals for light

emitting diodes using the finite difference time domain technique, Proceedings of SPIE, Vol 3283, (1998) pp.184-190, ISSN 0277-786X

Cohn, S., B (1947) Properties of Ridge Wave Guide, Proceedings of the IRE, Vol.35, (Aug

1947) pp 783-788, ISSN 0096-8390

Kokubo, Y.; Maki, D & Kawai, T (2007a) Dual-Band Metallic Waveguide with Low

Dielectric Constant Material, 37th European Microwave Conference Proceedings,

pp.890-892, ISBN 978-2-87487-000-2, Munich, Germany, Oct 2007, EuMA, Belgium Kokubo, Y.; Yoshida, S & Kawai, T (2007b) Economic Setup for a Dual-band Metallic

Waveguide with Dual In-line Dielectric Rods, IEICE Transactions on Electronics,

Vol.E90-C, No.12, (Dec 2007) pp.2261-2262, ISSN 0916-8524

Kokubo, Y & Kawai, T (2008) A Frequency Multiplexer/Demultiplexer for Dual Frequency

Waveguide, 38th European Microwave Conference Proceedings, (Oct 2008) pp.24-27,

ISBN 978-2-87487-005-7, Amsterdam, The Netherland, Oct 2008, EuMA, Belgium Maradudin, A A & McGurn, A R (1993) Photonic band structure of a truncated, two-

dimensional, periodic dielectric medium, Journal of the Optical Society of America B,

Vol.10, No.2, (1993) pp 307-313, ISSN 0740-3224

Shibano, T.; Maki, D & Kokubo, Y (2006) Dual Band Metallic Waveguide with Dual in-line

Dielectric Rods, IEICE Transactions on Electronics, Vol.J89-C, No.10, (Oct 2006)

pp.743-744, ISSN 1345-2827 (Japanese Edition) ; Correction and supplement, ibid, Vol.J90-C, No.3, (Mar 2007) p.298, ISSN 1345-2827 (Japanese Edition)

Tayeb, G & Maystre, D (1997) Rigorous theoretical study of finite-size two-dimensional

photonic crystals doped by microcavities, Journal of the Optical Society of America A,

Vol 14, No.12, (Dec 1997) pp 3323-3332, ISSN 1084-7529

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Applications of On-Chip Coplanar

Waveguides to Design Local

Oscillators for Wireless Communications System

Ramesh K Pokharel, Haruichi Kanaya

and Keiji Yoshida

Kyushu University

Japan

1 Introduction

On-chip distributed transmission line resonators in CMOS technology have become the

interest of research subjects recently (Ono et al 2001; Umeda et al., 1994; Kanaya et al., 2006;

Wolf, 2006) because of their size which becomes more compact, as the frequency of

application increases Among the various transmission lines, coplanar waveguide (CPW)

has more engineering applications (Toyoda, 1996; Civello, 2005) because it is easy to

fabricate by LSI technology since the signal line and ground plane exist on the same plane so

that no via holes are required for integrating active components such as transistors on

Si-substrate (Toyoda, 1996)

The applications of the CPW were reported for many on-chip LSI components The CPW

was exploited as an inductor and used to design a conventional-type matching circuit for

LNA (Ono et al., 2001) in microwave-band frequency, and they are most popular in

monolithic microwave integrated circuit (MMIC) (Umeda et al., 1994) However, the

application of CPW lines as an inductor takes larger space than the conventional spiral

inductors (Umeda et al., 1994) Some of the present authors have also implemented the

on-chip CPW impedance-matching circuit for a 2.4 GHz RF front-end (Kanaya et al., 2006) and

for 5GHz band power amplifier (Pokharel et al., 2008) using impedance inverters In

designing the matching circuits using impedance inverters and quarter wavelength

resonators realized by on-chip CPW (Kanaya et al, 2006; Pokharel et al., 2008) the size of the

matching circuits becomes compact thus reducing the chip area by about 30% than using

spiral inductors for 2.4GHz-band applications and 40% for 5 GHz-band applications

However, the applications of on-chip CPW resonators in designing other components such

as a voltage-controlled oscillator (VCO) have not been reported yet A conventional VCO

consists of a LC-resonator to produce an oscillation at the frequency band of interest, and

this LC-resonator may be replaced by a CPW resonator Such possibilities are investigated in

this paper In a conventional VCO, the performance such as phase noise of the VCO

depends on the quality (Q) factor of the LC resonator Usually, a spiral inductor is used in

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the resonator and these have quite low Q’s of around 3-5 at GHz frequency range and on the

other hand, it takes large on-chip area in the expensive silicon substrate The inductor can be

either resonated with the device drain capacitance or by adding a shunt capacitor (on chip

or off) Using bond wires instead of on-chip spiral inductors allows the design of low phase

noise oscillators but makes the fabrication more difficult as it is difficult to precisely set the

length of the bond wire Also for use in Phase Locked Loop (PLL) applications it is

necessary to have variable frequency or so called higher frequency tuning range (FTR)

Therefore, it is not a wise practice to use bond wires in designing a VCO due to design

difficulties in estimating the bond wires inductances

In this paper, first, we propose a design method of a VCO using on-chip CPW resonator

thus replacing an LC-resonator First, transmission characteristics of the on-chip meander

CPW resonator fabricated using TSMC 0.18 m CMOS technology are investigated

experimentally and an equivalent circuit is developed Later, the application of on-chip

resonator is also demonstrated to design 10 bits digitally-controlled oscillator (DCO) The

derived equivalent circuit is used to carry out the post-layout simulation of the chip One of

the advantages of the proposed method to design VCO and DCO using on-chip CPW

resonator than using a LC-resonator is smaller chip area

2 Design of On-Chip CPW Resonator and Its Equivalent Circuits

In this paper, we use Advanced Design System (ADS2008A, Agilent Technologies) for

designing active elements and Momentum (Agilent Technologies) for passive elements for

schematic design Co-simulation option was used for electromagnetic characterization of

hybrid structures consisting of active and passive elements together We first develop the

equivalent circuit for on-chip meander CPW resonator using experimental results and latter,

the circuit is used to carry out the post-layout simulation of the chip

The on-chip meander CPW resonator is designed, fabricated, and measured using TSMC

0.18 m CMOS technology This process has 1-poly and 6-metal layers and the thickness of

the top metal is 3.1 m The conductance of the metal and dielectric permittivity (r) of the

SiO2 are 4.1x107 S/m and 4.1, respectively The upper layer is covered by lamination whose

relative permittivity is 7.9

Fig 1 shows the layout and chip photos of on-chip CPW resonator designed and

characterized by EM simulator In Fig 1(a), the enlarged portion of the layout is illustrated

to show its structure in detail where the signal line and slot size is 5 m each, respectively

Bottom metal (Metal-1) is used as ground plane covering all portion of CPW to reduce the

losses Therefore, we prefer to call this CPW as conductor-backed CPW Total length of the

resonator is 3300 m which is supposed to be shorter than a quarter-wavelength resonator at

5.2 GHz The chip photo of the on-chip CPW resonator is shown in Fig 1(b) and Fig 1(c)

Please note that a small stub at the center CPW pad (dummy pad of right side) in Fig 1(b) is

to de-embed the interconnect between metal 6 terminal of the CPW resonator and the pad

The microwave characteristics are measured by using air coplanar probes (Cascade

Microtech, GSG150) and vector network analyzer (HP, HP8722C) in Air coplanar probe

station (Cascade Microtech Inc.) The CPW pads are 100m square and have coplanar

configurations so that characteristic impedance is 50 

(a) Layout of CPW resonator showing enlarged section for illustration of its structure

(b) Dummy chip (c) On-chip CPW meander resonator Fig 1 Layouts and chip photographs of CPW resonator

The measured data must be de-embedded in order to remove the parasitic effects of interconnects, pads and contacts surrounding the device (Civello, 2005) Therefore, in Fig 1(b), chip photo of a dummy pad and in Fig 1(c), chip photo of the CPW resonator are shown In order to de-embed the measured raw data, at first, we measure S-parameters of total (Fig 1(c)) and open dummy chip (Fig 1(b)), respectively Next, S-parameters are transformed into Y-parameters according to Equation (1) to get the Y-parameters (YTML) of the transmission-line resonator only

(1)

Y-parameters are then converted to Z-parameters in order to compare the results between simulation using the Equivalent circuits of Fig 2 In Fig 2, two equivalent circuits are developed using 2-stages and 5-stages for CPW resonator in meander structure, where ideal transmission lines are represented by the parameters such as characteristic impedance (Z0), electrical length of each part (E), and frequency (F) Furthermore, C1 represents the mutual capacitance between the meander lines, R1 is the resistive loss of the line in each segment, and the parameters R (resistance), C (Capacitance) represent the silicon substrate of the corresponding segment In Fig 2(b), where 5-stage model of equivalent circuit is shown, the meander line is divided into shorter segments, therefore parameters of each segment of the model such as R1, C1, E will differ from 2-stage model of Fig 2(a) Each parameters in both models are noted below each figure Here, model parameters for Si-substrate (R, C) are

TML total dummy[ ]Y [ ]Y [ ]Y

Ground

GroundSignalGround

GroundSignal

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